Wireless power feeder, wireless power receiver, and wireless power transmission system

ABSTRACT

Power is fed from a feeding coil L 2  to a receiving coil L 3  by magnetic resonance. A VCO  202  alternately turns ON/OFF switching transistors Q 1  and Q 2  at a drive frequency fo, whereby AC power is fed to the feeding coil L 2 , and then the AC power is fed from the feeding coil L 2  to the receiving coil L 3 . A phase detection circuit  114  detects a phase difference between the current phase and voltage phase, and the VCO  202  adjusts the drive frequency fo such that the phase difference becomes zero. When load voltage is changed, the detected current phase value is adjusted with the result that the drive frequency fo is adjusted.

This is a continuation-in-part of application Ser. No. 12/944,566, filedon Nov. 11, 2010.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a wireless AC power feeding and, moreparticularly, to power control thereof.

2. Description of Related Art

A wireless power feeding technique of feeding power without a power cordis now attracting attention. The current wireless power feedingtechnique is roughly divided into three: (A) type utilizingelectromagnetic induction (for short range); (B) type utilizing radiowave (for long range); and (C) type utilizing resonance phenomenon ofmagnetic field (for intermediate range).

The type (A) utilizing electromagnetic induction has generally beenemployed in familiar home appliances such as an electric shaver;however, it can be effective only in a short range of severalcentimeters. The type (B) utilizing radio wave is available in a longrange; however, it cannot feed big electric power. The type (C)utilizing resonance phenomenon is a comparatively new technique and isof particular interest because of its high power transmission efficiencyeven in an intermediate range of about several meters. For example, aplan is being studied in which a receiving coil is buried in a lowerportion of an EV (Electric Vehicle) so as to feed power from a feedingcoil in the ground in a non-contact manner. The wireless configurationallows a completely insulated system to be achieved, which is especiallyeffective for power feeding in the rain. Hereinafter, the type (C) isreferred to as “magnetic field resonance type”.

The magnetic field resonance type is based on a theory published byMassachusetts Institute of Technology in 2006 (refer to Patent Document1). In Patent Document 1, four coils are prepared. The four coils arereferred to as “exciting coil”, “feeding coil”, “receiving coil”, and“loading coil” in the order starting from the feeding side. The excitingcoil and feeding coil closely face each other for electromagneticcoupling. Similarly, the receiving coil and loading coil closely faceeach other for electromagnetic coupling. The distance (intermediatedistance) between the feeding coil and receiving coil is larger than thedistance between the exciting coil and feeding coil and distance betweenthe receiving coil and loading coil. This system aims to feed power fromthe feeding coil to receiving coil.

When AC power is fed to the exciting coil, current also flows in thefeeding coil according to the principle of electromagnetic induction.When the feeding coil generates a magnetic field to cause the feedingcoil and receiving coil to magnetically resonate, large current flows inthe receiving coil. At this time, current also flows in the loading coilaccording to the principle of electromagnetic induction, and power istaken out from a load connected in series to the loading coil. Byutilizing the magnetic field resonance phenomenon, high powertransmission efficiency can be achieved even if the feeding coil andreceiving coil are largely spaced from each other.

CITATION LIST Patent Document

-   [Patent Document 1] U.S. Pat. Appln. Publication No. 2008/0278264-   [Patent Document 2] Jpn. Pat. Appln. Laid-Open Publication No.    2006-230032-   [Patent Document 3] International Publication Pamphlet No.    WO2006/022365-   [Patent Document 4] U.S. Pat. Appln. Publication No. 2009/0072629-   [Patent Document 5] U.S. Pat. Appln. Publication No. 2009/0015075-   [Patent Document 6] Jpn. Pat. Appln. Laid-Open Publication No.    2008-172872-   [Patent Document 7] Jpn. Pat. Appln. Laid-Open Publication No.    2006-74848-   [Patent Document 8] Jpn. Pat. Appln. Laid-Open Publication No.    2003-33011

The present inventor considers that a mechanism for automaticallycontrolling feeding power so as to make output power stable is requiredin order to extend the applicability of wireless power feeding. In anon-contact type power feeder disclosed in Patent Document 7, which isof the type (A), a secondary side unit on the receiving side notifies aprimary side unit on the transmission side of the magnitude of outputvoltage, and the primary side unit controls feeding power in accordancewith the output voltage. More specifically, a signal indicating themagnitude of the output voltage is transmitted from a coil L4 (secondaryside unit) to a coil L3 (primary side unit).

In the non-contact type power feeder of Patent Document 7, it issilently assumed that the resonance frequency of a primary side seriesresonance circuit or a secondary side series resonance circuit is set toa fixed value. However, in the case of power feeding of a magnetic fieldresonance type, the resonance frequency is liable to change depending onthe positional relationship between feeding and receiving coils and,therefore, the mechanism of Patent Document 7 cannot practically beapplied to the magnetic field resonance type. Further, it is consideredthat, in the case of the magnetic field resonance type, a magnetic fieldgenerated by feeding or receiving coil greatly affects on signaltransmission from the coil L4 to coil L3 made by using electromagneticwave.

SUMMARY

A main object of the present invention is to effectively control feedingpower in wireless power feeding of a magnetic field resonance type.

A wireless power feeder according to a first aspect of the presentinvention is a device that feeds power from a feeding coil to areceiving coil by wireless using a magnetic field resonance phenomenonbetween the feeding coil and receiving coil. The wireless power feederincludes: a feeding coil circuit that includes the feeding coil; a powertransmission control circuit that feeds AC power to the feeding coil ata drive frequency; a phase detection circuit that detects a phasedifference between the voltage phase and current phase of the AC power;and a signal receiving circuit that receives an output signal indicatingthe magnitude of an output from the power receiving side. The powertransmission control circuit adjusts the drive frequency so as to reducethe phase difference. The phase detection circuit performs ex-postadjustment of the detected value of both or one of the voltage andcurrent phases according to the output signal.

The current phase and voltage phase of the AC power are compared todetect the phase difference between the current and voltage phases.Adjusting the drive frequency so as to reduce the detected phasedifference allows the drive frequency to track the resonance frequency.As a result, even if the resonance frequency is changed, the powertransmission efficiency is easily kept constant. Further, ex-postadjustment of the voltage phase or current phase in accordance with avariation of the output voltage, even if it occurs, causes the drivefrequency to be changed in accordance with the adjusted phasedifference. Thus, the feeding power can be feedback-controlled using thedrive frequency as a parameter, making it easy to stabilize the outputvoltage.

The phase detection circuit may convert both or one of voltage andcurrent components of the AC power into a signal having a saw-toothwaveform for detection of the phase difference. The signal receivingcircuit may receive the output signal as a light signal such as aninfrared ray. The output signal may be an AC signal indicating themagnitude of the output by the magnitude of signal frequency. The phasedetection circuit may compare a first phase value indicating a timing atwhich the voltage level of the AC power becomes a first reference valueand a second phase value indicating a timing at which the current levelof the AC power becomes a second reference value to detect the phasedifference and change both or one of the first and second referencevalues based on the output signal to perform ex-post adjustment of bothor one of the first and second phase values.

The wireless power feeder may further include an exciting coil that ismagnetically coupled to the feeding coil and feeds AC power fed from thepower transmission control circuit to the feeding coil. The powertransmission control circuit may include first and second current pathsand make first and second switches connected in series respectively tothe first and second current paths alternately conductive at the drivefrequency to feed the AC power to the exciting coil.

The wireless power feeder may further include a detection coil thatgenerates inductive current using a magnetic field generated by the ACpower. The phase detection circuit may measure the phase of theinductive current flowing in the detection coil to achieve measurementof the current phase of the AC power. Since the current phase ismeasured from the inductive current flowing in the detection coil, ameasurement load is difficult to be directly applied to the feedingcoil. The detection coil may generate the inductive current using amagnetic field generated by the AC power flowing in the feeding coil.

The power supply circuit may make the feeding coil that does notsubstantially resonate with circuit elements on the power feeding sidefeed the AC power to the receiving coil. The “does not substantiallyresonate” mentioned here means that the resonance of the feeding coil isnot essential for the wireless power feeding, but does not mean thateven an accidental resonance of the feeding coil with some circuitelement is eliminated. A configuration may be possible in which thefeeding coil does not form, together with circuit elements on the powerfeeding side, a resonance circuit having a resonance point correspondingto the resonance frequency of the receiving coil. Further, aconfiguration may be possible in which no capacitor is connected inseries or in parallel to the feeding coil.

The feeding coil circuit may be constructed as a circuit that resonatesat the resonance frequency of the receiving coil.

A wireless power receiver according to a second aspect of the presentinvention is a device that receives AC power fed from the wireless powerfeeder as described above by wireless at a receiving coil. The wirelesspower receiver includes: a receiving coil circuit that includes thereceiving coil and a capacitor; a loading circuit that includes aloading coil that is magnetically coupled to the receiving coil toreceive the AC power from the receiving coil and a load that receivespower from the loading coil; and a signal transmission circuit thattransmits, to the wireless power feeder, an output signal indicating themagnitude of output voltage to be applied to a part of the loadingcircuit.

The signal transmission circuit may transmit the output signal as asignal indicating a difference value between the output voltage and areference voltage. The value of the reference voltage may manually beadjustable. The output signal may be an AC signal indicating themagnitude of the output voltage by the magnitude of signal frequency.The output voltage may be generated as a DC current by the rectificationcircuit provided in the loading circuit.

The receiving coil circuit may be constructed as a circuit thatresonates at the resonance frequency of the feeding coil.

A wireless power transmission system according to a third aspect of thepresent invention is a system that feeds power from a feeding coil to areceiving coil by wireless using a magnetic field resonance phenomenonbetween the feeding coil and receiving coil. The system includes: afeeding coil circuit that includes the feeding coil; a powertransmission control circuit that feeds AC power to the feeding coil ata drive frequency; a receiving coil circuit that includes the receivingcoil and a capacitor; a loading circuit that includes a loading coilthat is magnetically coupled to the receiving coil to receive the ACpower from the receiving coil and a load that receives power from theloading coil; and a phase detection circuit that detects a phasedifference between the voltage phase and current phase of the AC power.The power transmission control circuit adjusts the drive frequency so asto reduce the phase difference. The phase detection circuit performsex-post adjustment of the detected value of both or one of the voltageand current phases according to the magnitude of an output voltage to beapplied to a part of the loading circuit.

A wireless power feeder according to a fourth aspect of the presentinvention is a device that feeds power from a feeding coil to areceiving coil by wireless using a magnetic field resonance phenomenonbetween the feeding coil and receiving coil. The wireless power feederincludes: a feeding coil circuit that includes the feeding coil; a powertransmission control circuit that feeds AC power to the feeding coil ata drive frequency; a phase detection circuit that detects a phasedifference between the voltage phase and current phase of the AC power;and a signal receiving circuit that receives an output signal indicatingan output by a duty ratio from the power receiving side of the AC powerand DC-converts the output signal in accordance with the duty ratio. Thepower transmission control circuit adjusts the drive frequency so as toreduce the phase difference. The phase detection circuit performsex-post adjustment of the detected value of both or one of the voltageand current phases in accordance with the signal level of theDC-converted output signal.

The current phase and voltage phase of the AC power are compared todetect the phase difference between the current and voltage phases.Adjusting the drive frequency so as to reduce the detected phasedifference allows the drive frequency to track the resonance frequency.As a result, even if the resonance frequency is changed, the powertransmission efficiency is easily kept constant. Further, ex-postadjustment of the voltage phase or current phase in accordance with avariation of the output voltage, even if it occurs, causes the drivefrequency to be changed in accordance with the adjusted phasedifference. Thus, the feeding power can be feedback-controlled using thedrive frequency as a parameter, making it easy to stabilize the outputvoltage.

The phase detection circuit may compare a first phase value indicating atiming at which the voltage level of the AC power becomes a firstreference value and a second phase value indicating a timing at whichthe current level of the AC power becomes a second reference value todetect the phase difference and change both or one of the first andsecond reference values in accordance with the signal level to performex-post adjustment of both or one of the first and second phase values.The signal receiving circuit may receive the output signal as a lightsignal such as an infrared ray.

The wireless power feeder may further include an exciting coil that ismagnetically coupled to the feeding coil and feeds AC power fed from thepower transmission control circuit to the feeding coil. The powertransmission control circuit may include first and second current pathsand make first and second switches connected in series respectively tothe first and second current paths alternately conductive at the drivefrequency to feed the AC power to the exciting coil.

The wireless power feeder may further include a detection coil thatgenerates inductive current using a magnetic field generated by the ACpower. The phase detection circuit may measure the phase of theinductive current flowing in the detection coil to achieve measurementof the current phase of the AC power. Since the current phase ismeasured from the inductive current flowing in the detection coil, ameasurement load is difficult to be directly applied to the feedingcoil. The detection coil may generate the inductive current using amagnetic field generated by the AC power flowing in the feeding coil.

A wireless power receiver according to a fifth aspect of the presentinvention is a device that receives AC power fed from the wireless powerfeeder as described above by wireless at a receiving coil. The wirelesspower receiver includes: a receiving coil circuit that includes thereceiving coil and a capacitor; a loading circuit that includes aloading coil that is magnetically coupled to the receiving coil toreceive the AC power from the receiving coil and a load that receivespower from the loading coil; and a signal transmission circuit thattransmits, to the wireless power feeder, an output signal indicatingoutput voltage to be applied to a part of the loading circuit by a dutyratio.

The signal transmission circuit may transmit the output signal as asignal indicating a difference value between the output voltage and areference voltage by the duty ratio. The value of the reference voltagemay manually be adjustable.

The wireless power receiver may further include: a control signalgeneration circuit that generates a control signal at a predeterminedcontrol frequency; and a comparison circuit that generates an enablesignal when a predetermined magnitude relationship is establishedbetween the signal level of the control signal and the output voltage.The signal transmission circuit may determine the duty ratio of theoutput signal based on the duty ratio of the enable signal. The“predetermined magnitude relationship” may be that the level of thecontrol signal is higher than a predetermined level representing avariation of the output voltage, or vice versa.

The wireless power receiver may further include a reference signalgeneration circuit that generates a reference signal having a referencefrequency higher than the control frequency. The signal transmissioncircuit may transmit the reference signal as the output signal onlywhile the enable signal is being generated.

A wireless power transmission system according to a sixth aspect of thepresent invention is a system that feeds power from a feeding coil to areceiving coil by wireless using a magnetic field resonance phenomenonbetween the feeding coil and receiving coil. The system includes: apower transmission control circuit that feeds AC power to the feedingcoil at a drive frequency; a feeding coil circuit that includes thefeeding coil; a receiving coil circuit that includes the receiving coiland a capacitor; a loading circuit that includes a loading coil that ismagnetically coupled to the receiving coil to receive the AC power fromthe receiving coil and a load that receives power from the loading coil;a phase detection circuit that detects a phase difference between thevoltage phase and current phase of the AC power; a signal transmissioncircuit that transmits, to the power feeder side, an output signalindicating output voltage to be applied to a part of the loading circuitby a duty ratio; and a signal receiving circuit that receives the outputsignal at the power feeder side and DC-converts the output signal inaccordance with the duty ratio. The power transmission control circuitadjusts the drive frequency so as to reduce the phase difference. Thephase detection circuit performs ex-post adjustment of the detectedvalue of both or one of the voltage and current phases in accordancewith the signal level of the DC-converted output signal.

It is to be noted that any arbitrary combination of the above-describedstructural components and expressions changed between a method, anapparatus, a system, etc. are all effective as and encompassed by thepresent embodiments.

According to the present invention, it is possible to effectivelycontrol transmission power in wireless power feeding of a magnetic fieldresonance type.

BRIEF DESCRIPTION OF THE DRAWINGS

The above features and advantages of the present invention will be moreapparent from the following description of certain preferred embodimentstaken in conjunction with the accompanying drawings, in which:

FIG. 1 is a view illustrating operation principle of a wireless powertransmission system according to first and second embodiments;

FIG. 2 is a system configuration view of a wireless power transmissionsystem in a first embodiment;

FIG. 3 is a graph illustrating a relationship between load current andload voltage;

FIG. 4 is a graph illustrating inter-coil distance and load voltage;

FIG. 5 is a graph illustrating a relationship between the impedance of afeeding coil circuit and drive frequency;

FIG. 6 is a graph illustrating a relationship between output powerefficiency and drive frequency;

FIG. 7 is a circuit diagram of a signal transmission circuit in thefirst embodiment;

FIG. 8 is a graph illustrating a relationship between the signalfrequency in a VF converter and a T0 signal;

FIG. 9 is a circuit diagram of a signal receiving circuit in the firstembodiment;

FIG. 10 is a graph illustrating a relationship between signal frequencyin an FV converter and a T2 signal;

FIG. 11 is a circuit diagram of a current phase detection circuit in thefirst embodiment;

FIG. 12 is a time chart illustrating a relationship between S0 and S1signals;

FIG. 13 is a circuit diagram illustrating a modification example of awireless power receiver in the first embodiment;

FIG. 14 is a system configuration view illustrating a modificationexample of the wireless power transmission system in the firstembodiment;

FIG. 15 is a system configuration view of a wireless power transmissionsystem in a second embodiment;

FIG. 16 is a graph illustrating a relationship between load current andload voltage;

FIG. 17 is a graph illustrating inter-coil distance and load voltage;

FIG. 18 is a graph illustrating a relationship between the impedance ofa feeding coil circuit and drive frequency;

FIG. 19 is a graph illustrating a relationship between output powerefficiency and drive frequency;

FIG. 20 is a circuit diagram of a control signal generation circuit inthe second embodiment;

FIG. 21 is a time chart illustrating a relationship among T0, T2 and T4signals;

FIG. 22 is a circuit diagram of a signal transmission circuit in thesecond embodiment;

FIG. 23 is a time chart illustrating a relationship among T2, T3, and T6signals;

FIG. 24 is a circuit diagram illustrating a current phase detectioncircuit and a signal receiving circuit in the second embodiment;

FIG. 25 is a time chart illustrating a relationship among S1, S3, and T5signals;

FIG. 26 is a time chart illustrating a relationship between S1 and S2signals;

FIG. 27 is a circuit diagram illustrating a modification example of awireless power receiver in the second embodiment;

FIG. 28 is a system configuration view illustrating a modificationexample of the wireless power transmission system in the secondembodiment;

FIG. 29 is a view illustrating operation principle of the wireless powertransmission system according to third and fourth embodiments;

FIG. 30 is a system configuration view of the wireless powertransmission system according to the third embodiment; and

FIG. 31 is a system configuration view of the wireless powertransmission system according to the fourth embodiment.

DETAILED DESCRIPTION OF THE EMBODIMENTS

A preferred embodiment of the present invention will be described belowwith reference to the accompanying drawings.

FIG. 1 is a view illustrating operation principle of a wireless powertransmission system 100 according to the first and second embodiment.The wireless power transmission system 100 in the first embodimentincludes a wireless power feeder 116 and a wireless power receiver 118.The wireless power feeder 116 includes a power feeding LC resonancecircuit 300. The wireless power receiver 118 includes a receiving coilcircuit 130 and a loading circuit 140. A power receiving LC resonancecircuit 302 is formed by the receiving coil circuit 130.

The power feeding LC resonance circuit 300 includes a capacitor C2 and afeeding coil L2. The power receiving LC resonance circuit 302 includes acapacitor C3 and a receiving coil L3. The values of the capacitor C2,feeding coil L2, capacitor C3, and receiving coil L3 are set such thatthe resonance frequencies of the feeding LC resonance circuit 300 andreceiving LC resonance circuit 302 coincide with each other in a statewhere the feeding coil L2 and receiving coil L3 are disposed away fromeach other far enough to ignore the magnetic field couplingtherebetween. This common resonance frequency is assumed to be fr0.

In a state where the feeding coil L2 and receiving coil L3 are broughtclose to each other in such a degree that they can bemagnetic-field-coupled to each other, a new resonance circuit is formedby the power feeding LC resonance circuit 300, power receiving LCresonance circuit 302, and mutual inductance generated between them. Thenew resonance circuit has two resonance frequencies fr1 and fr2(fr1<fr0<fr2) due to the influence of the mutual inductance. When thewireless power feeder 116 supplies AC power from a power feeding sourceVG to the power feeding LC resonance circuit 300 at the resonancefrequency fr1, the power feeding LC resonance circuit 300 constitutingapart of the new resonance circuit resonates at a resonance point 1(resonance frequency fr1). When the power feeding LC resonance circuit300 resonates, the feeding coil L2 generates an AC magnetic field of theresonance frequency fr1. The power receiving LC resonance circuit 302constituting apart of the new resonance circuit also resonates byreceiving the AC magnetic field. When the power feeding LC resonancecircuit 300 and power receiving LC resonance circuit 302 resonate at thesame resonance frequency fr1, wireless power feeding from the feedingcoil L2 to receiving coil L3 is performed with the maximum powertransmission efficiency. Received power is taken from a load LD of thewireless power receiver 118 as output power. Note that the new resonancecircuit can resonate not only at the resonance point 1 (resonancefrequency fr1) but also at a resonance point 2 (resonance frequencyfr2).

Although FIG. 1 illustrates a configuration in which the wireless powerfeeder 116 does not include an exciting coil L1, the basic operationprinciple of the wireless power feeder 116 is the same as in the casewhere the wireless power feeder 116 includes the exciting coil L1. Aconfiguration in which the wireless power feeder 116 does not includethe exciting coil L1 will be described later using FIG. 14 and the like.

First Embodiment

FIG. 2 is a system configuration view of a wireless power transmissionsystem 100 according to the first and second embodiment. The wirelesspower transmission system 100 includes a wireless power feeder 116 and awireless power receiver 118. The wireless power feeder 116 includes, asbasic components, a power transmission control circuit 200, an excitingcircuit 110, a feeding coil circuit 120, a phase detection circuit 114,and a signal receiving circuit 112. The wireless power receiver 118includes a receiving coil circuit 130, a loading circuit 140, and asignal transmission circuit 122.

A distance (hereinafter, referred to as “inter-coil distance”) of about0.2 m to 1.0 m is provided between a feeding coil L2 of the feeding coilcircuit 120 and a receiving coil L3 of the receiving coil circuit 130.The wireless power transmission system 100 mainly aims to feed powerfrom the feeding coil L2 to receiving coil L3 by wireless. In the firstembodiment, a description will be made assuming that resonance frequencyfr1 is 100 kHz. The wireless power transmission system of the firstembodiment may be made to operate in a high-frequency band like ISM(Industry-Science-Medical) frequency band. A low frequency band isadvantageous over a high frequency band in reduction of cost of aswitching transistor (to be described later) and reduction of switchingloss. In addition, the low frequency band is less constrained by RadioAct.

The exciting circuit 110 is a circuit in which an exciting coil L1 and atransformer TS2 secondary coil Li are connected in series. Thetransformer TS2 secondary coil Li constitutes a coupling transformer TS2together with a transformer TS2 primary coil Lb and receives AC powerfrom the power transmission control circuit 200 by electromagneticinduction. The number of windings of the exciting coil L1 is 1, diameterof a conductive wire is 5 mm, and shape of the exciting coil L1 itselfis a square of 210 mm×210 mm. In FIG. 2, the exciting coil L1 isrepresented by a circle for clarification. Other coils are alsorepresented by circles for the same reason. All the coils illustrated inFIG. 2 are made of copper. Current I1 flowing in the exciting circuit110 is AC.

The feeding coil circuit 120 is a circuit in which a feeding coil L2 anda capacitor C2 are connected in series. The exciting coil L1 and feedingcoil L2 face each other. The distance between the exciting coil L1 andfeeding coil L2 is as comparatively small as 10 mm or less. Thus, theexciting coil L1 and feeding coil L2 are electromagnetically stronglycoupled to each other. The number of windings of the feeding coil L2 is7, diameter of a conductive wire is 5 mm, and shape of the feeding coilL2 itself is a square of 280 mm×280 mm. When the AC current I1 is madeto flow in the exciting coil L1, an electromotive force occurs in thefeeding coil L2 according to the principle of electromagnetic inductionto cause AC current I2 to flow in the feeding coil circuit 120. The ACcurrent I2 is considerably larger than the AC current I1. The values ofthe feeding coil L2 and capacitor C2 are set such that the resonancefrequency fr1 of the feeding coil circuit 120 is 100 kHz.

The receiving coil circuit 130 is a circuit in which a receiving coil L3and a capacitor C3 are connected in series. The feeding coil L2 andreceiving coil L3 face each other. The number of windings of thereceiving coil L3 is 7, diameter of a conductive wire is 5 mm, and shapeof the receiving coil L3 itself is a square of 280 mm×280 mm. The valuesof the receiving coil L3 and capacitor C3 are set such that theresonance frequency fr1 of the receiving coil circuit 130 is also 100kHz. Thus, the feeding coil L2 and receiving coil L3 need not have thesame shape. When the feeding coil L2 generates a magnetic field at theresonance frequency fr1=100 kHz, the feeding coil L2 and receiving coilL3 magnetically resonate, causing large current I3 to flow in thereceiving coil circuit 130.

The loading circuit 140 is a circuit in which a loading coil L4 isconnected to a load LD through a rectification circuit 124 and ameasurement circuit 126. The receiving coil L3 and loading coil L4 faceeach other. The distance between the receiving coil L3 and loading coilL4 is as comparatively small as about 10 mm or less. Thus, the receivingcoil L3 and loading coil L4 are electromagnetically strongly coupled toeach other. The number of windings of the loading coil L4 is 1, diameterof the wire of the loading coil L4 is 5 mm, and shape of the loadingcoil L4 itself is a square of 300 mm×300 mm. When the current I3 is madeto flow in the receiving coil L3, an electromotive force occurs in theloading circuit 140 to cause AC current I4 to flow in the loadingcircuit 140. The AC current I4 is rectified into DC current by therectification circuit 124. Although part of the DC current flows in themeasurement circuit 126, most of the DC current flows in the load LD asDC current I5. The rectification circuit 124 is a general circuitconstituted by a bridge circuit 128 and a capacitor C5. The details ofthe measurement circuit 126 will be described later.

The AC power fed from the feeding coil L2 of the wireless power feeder116 is received by the receiving coil L3 of the wireless power receiver118 and then extracted from the load LD as DC power. Voltage applied tothe load LD is referred to as “load voltage V5”.

If the load LD is connected in series to the receiving coil circuit 130,the Q-value of the receiving coil circuit 130 is degraded. Therefore,the receiving coil circuit 130 for power reception and loading circuit140 for power extraction are separated from each other. In order toenhance the power transmission efficiency, the center lines of thefeeding coil L2, receiving coil L3, and loading coil L4 are preferablymade to coincide with one another.

The measurement circuit 126 includes resistors R1 and R2, a controlpower supply Vs, and a comparator 132. The load voltage V5 is divided bythe resistors R1 and R2. Voltage applied to both ends of the resistorsR2 is referred to as “output voltage”. The potential at a connectingpoint T between the resistors R1 and R2 is input to the negativeterminal of the comparator 132 as “measurement potential”. A controlpower supply Vs is connected to the positive terminal of the comparator132. Input voltage at the positive terminal of the comparator 132generated by the control power supply Vs is referred to as “referencepotential”. The control power supply Vs is a variable DC voltage supply,and the voltage thereof can arbitrarily be adjusted. The comparator 132amplifies a difference (hereinafter, referred to as “correctionvoltage”) between the measurement potential and reference potential andoutputs the amplified value as a T0 signal. The T0 signal is a DCvoltage signal and indicates the magnitude of the correction voltage. Inother words, the T0 signal is a signal indicating a change in the loadvoltage V5. Although the details will be described later, in thewireless power transmission system 100 of the first embodiment, thefeeding power is controlled so as to make the correction voltage be zeroto thereby stabilize the output voltage (load voltage V5).

The signal transmission circuit 122 converts the T0 signal as the DCvoltage signal into a T1 signal as an AC light signal. The T1 signal isan “output signal” indicating the magnitude of output and received bythe signal receiving circuit 112 of the wireless power feeder 116. Thepower feeding side can recognize the magnitude of the correction voltagebased from the T1 signal. A circuit configuration and processing of thesignal transmission circuit 122 will be described later using FIGS. 7and 8. A circuit configuration and processing of the signal receivingcircuit 112 will be described later using FIGS. 9 and 10.

A configuration of the power transmission control circuit 200 will bedescribed. A VCO (Voltage Controlled Oscillator) 202 is connected to theprimary side of the gate-drive transformer TS1. The VCO 202 functions asan “oscillator” that generates AC voltage Vo at the drive frequency fo.Although the waveform of the AC voltage Vo may be a sine wave, it isassumed here that the voltage waveform is a rectangular wave (digitalwave). The AC voltage Vo causes current to flow in a transformer TS1primary coil Lh alternately in both positive and negative directions. Atransformer TS1 primary coil Lh, a transformer TS1 secondary coil Lf,and a transformer TS1 secondary coil Lg constitute a gate-drive couplingtransformer TS1. Electromagnetic induction causes current to flow alsoin the transformer TS1 secondary coil Lf and transformer TS1 secondarycoil Lg alternately in both positive and negative directions.

As the VCO 202 in the first embodiment, a built-in unit (product serialnumber MC14046B) manufactured by Motorola, Inc is used. The VCO 202 alsohas a function of dynamically changing the drive frequency fo based onphase difference indicating voltage SC fed from the phase comparisoncircuit 150 (described later in detail).

The following description will be made assuming that the minimum valuefo1 of the drive frequency fo is 90 kHz, and the maximum value fo2thereof is 99 kHz. The appropriate range of the phase differenceindicating voltage SC is 1.0 (V) to 4.0 (V). The phase differenceindicating voltage SC and drive frequency fo are directly proportionalto each other. That is, when the phase difference indicating voltage SCis 1.0 (V), the drive frequency fo (=fo1) is 90 kHz, and when the phasedifference indicating voltage SC is 4.0 (V), the drive frequency fo(=fo2) is 99 kHz.

Capacitors CA and CB charged by a DC power supply Vdd each serve as apower supply for the power transmission control circuit 200. Thecapacitor CA is provided between points C and E of FIG. 2, and capacitorCB is provided between points E and D. Assuming that the voltage(voltage between points C and E) of the capacitor CA is VA, voltage(voltage between points E and D) of the capacitor CB is VB, VA+VB(voltage between points C and D) represents input voltage. That is, thecapacitors CA and CB each function as a DC voltage supply.

One end of the transformer TS1 secondary coil Lf is connected to thegate of a switching transistor Q1, and the other end of the transformerTS1 secondary coil Lf is connected to the source of a switchingtransistor Q1. One end of the transformer TS1 secondary coil Lg isconnected to the gate of a switching transistor Q2, and the other end ofthe transformer TS1 secondary coil Lg is connected to the source of aswitching transistor Q2. When VCO 202 generates AC voltage Vo at drivefrequency fo, voltage Vx (Vx>0) is alternately applied, at drivefrequency fo, to the gates of the switching transistors Q1 and Q2. As aresult, the switching transistors Q1 and Q2 are alternately turnedon/off at the drive frequency fo. The switching transistors Q1 and Q2are enhancement type MOSFET (Metal Oxide Semiconductor Field effecttransistor) having the same characteristics but may be other transistorssuch as a bipolar transistor. Further, other switches such as a relayswitch may be used in place of the transistor.

The drain of the switching transistor Q1 is connected to the positiveelectrode of the capacitor CA. The negative electrode of the capacitorCA is connected to the source of the switching transistor Q1 through thetransformer TS2 primary coil Lb. The source of the switching transistorQ2 is connected to the negative electrode of the capacitor CB. Thepositive electrode of the capacitor CB is connected to the drain of theswitching transistor Q2 through the transformer TS2 primary coil Lb.

Voltage between the source and drain of the switching transistor Q1 isreferred to as source-drain voltage VDS1, and voltage between the sourceand drain of the switching transistor Q2 is referred to as source-drainvoltage VDS2. Current flowing between the source and drain of theswitching transistor Q1 is referred to as source-drain current IDS1, andcurrent flowing between the source and drain of the switching transistorQ2 is referred to as source-drain current IDS2. The directions of arrowsin the diagram indicate the positive directions, and directions oppositeto the directions of the arrows indicate the negative directions.

When the switching transistor Q1 is turned conductive (ON), theswitching transistor Q2 is turned non-conductive (OFF). A main currentpath (hereinafter, referred to as “first current path”) at this timeextends from the positive electrode of the capacitor CA, passes throughthe point C, switching transistor Q1, transformer TS2 primary coil Lb,and point E in this order, and returns to the negative electrode of thecapacitor CA. The switching transistor Q1 functions as a switch forcontrolling conduction/non-conduction of the first current path.

When the switching transistor Q2 is turned conductive (ON), theswitching transistor Q1 is turned non-conductive (OFF). A main currentpath (hereinafter, referred to as “second current path”) at this timeextends from the positive electrode of the capacitor CB, passes throughthe point E, transformer TS2 primary coil Lb, switching transistor Q2,and point D in this order, and returns to the negative electrode of thecapacitor CB. The switching transistor Q2 functions as a switch forcontrolling conduction/non-conduction of the second current path.

Current flowing in the transformer TS2 primary coil Lb in the powertransmission control circuit 200 is referred to as “current IS”. Thecurrent IS is AC current, and the current flow in a first current pathis defined as the positive direction and current flow in a secondcurrent path is defined as the negative direction.

When the VCO 202 feeds the AC voltage Vo at the drive frequency fo, thefirst and second current paths are switched at the drive frequency fo.Since the AC current Is of the drive frequency fo flows in thetransformer TS2 primary coil Lb, the AC current I1 flows in the excitingcircuit 110 at the drive frequency fo, and the AC current I2 of thedrive frequency fo flows in the feeding circuit 120. The closer thevalue of the drive frequency fo is to the resonance frequency fr1, thehigher the power transmission efficiency becomes. When the drivefrequency fo is equal to the resonance frequency fr1, the feeding coilL2 of the feeding coil circuit 120 and capacitor C2 are in a resonancestate. The receiving coil circuit 130 is also a resonance circuit of theresonance frequency fr1, so that the feeding coil L2 and receiving coilL3 magnetically resonate. At this time, the maximum transmissionefficiency can be obtained.

In the case of the first embodiment, however, the resonance frequencyfr1 is not included in the operating range of the drive frequency fo, sothat the power transmission efficiency does not reach the maximum value.This is because priority is given to the stability of the load voltageV5 over the maximization of the power transmission efficiency. A changein the load voltage V5 can be detected from the correction voltage, sothat the wireless power feeder 116 automatically adjusts the drivefrequency fo so as to make the correction voltage be zero. The detailswill be described later.

The resonance frequency fr1 slightly changes depending on use conditionor use environment of the feeding coil circuit 120 or receiving coilcircuit 130. Further, in the case where the feeding coil circuit 120 orreceiving coil circuit 130 is replaced with new one, the resonancefrequency fr1 changes. Alternatively, there may be case where theresonance frequency fr1 needs to be changed aggressively by setting theelectrostatic capacitance of the capacitor C2 or capacitor C3 variable.Further, according to the experiment made by the present inventor, ithas been found that the resonance frequency starts falling when thedistance between the feeding coil L2 and receiving coil L3 is madesmaller to some extent. When the difference between the resonancefrequency fr1 and drive frequency fo changes, the power transmissionefficiency changes. When the power transmission efficiency changes, theload voltage V5 also changes. Therefore, in order to stabilize the loadvoltage V5, it is necessary to keep the difference between the resonancefrequency fr1 and the drive frequency fo constant even if the resonancefrequency fr1 changes.

The wireless power transmission system 100 in the first embodiment has adrive frequency tracking function of making the drive frequency foautomatically track a change of the resonance frequency fr1.

The phase detection circuit 114 includes a current phase detectioncircuit 144, a phase comparison circuit 150, and a low-pass filter 152.The low-pass filter 152 is a circuit in which a resistor R3 and acapacitor C6 are connected in series and cuts a high-frequency componentof the phase difference indicating voltage SC. As the phase comparisoncircuit 150 in the first embodiment, a built-in unit (Phase Comparator)(product serial number MC14046B) manufactured by Motorola is used, as inthe case of the VCO 202. Thus, the phase comparison circuit 150 and VCO202 can be implemented in one chip.

The current phase detection circuit 144 generates an S1 signal as asignal indicating a current phase. The S1 signal is input to the phasecomparison circuit 150. The AC voltage Vo generated by the VCO 202 isinput to the phase comparison circuit 150 as an S0 signal indicating avoltage phase. The phase comparison circuit 150 detects a deviation(phase difference) between the current phase and voltage phase from theS0 and S1 signals and generates the phase difference indicating voltageSC indicating the magnitude of the phase difference. Detecting the phasedifference allows detection of the magnitude of the deviation betweenthe resonance frequency fr1 and drive frequency fo. It is possible tokeep the phase difference between the drive frequency fo and theresonance frequency fr1 constant by controlling the drive frequency foaccording to the phase difference indicating voltage SC.

For example, when the drive frequency fo and resonance frequency fr1deviate from each other, the phase difference is accordingly increased,so that the phase comparison circuit 150 generates the phase differenceindicating voltage SC so as to reduce the phase difference. Thus, evenif the resonance frequency fr1 changes, it is possible to keep the powertransmission efficiency constant to thereby stabilize the load voltageV5. A circuit configuration of the current phase detection circuit 144will be described later using FIG. 11, and relationship between the S0and S1 signals will be described later using FIGS. 12 and 13.

The S0 signal may be obtained by connecting resistors to both ends ofthe transformer TS1 primary coil Lh in parallel to divide the AC voltageVo. Even in the case where the AC voltage Vo generated by the VCO 202 islarge, the AC voltage can be reduced to a manageable level by thevoltage division. The voltage phase may be measured from thesource-drain voltages VDS1 and VDS2 or source-gate voltages VGS1 andVGS2.

Even though the resonance frequency fr1 is constant, the load voltage V5may be changed in some cases. For example, in the case where the load LDis a variable resistor or in the case where the load LD is replaced withnew one, the load voltage V5 changes. In the first embodiment, a changein the load voltage V5 is detected as the correction voltage, and thedrive frequency fo is automatically adjusted so as to make thecorrection voltage be zero, whereby the load voltage V5 is stabilized.

The correction voltage is transmitted from the signal transmissioncircuit 122 to signal receiving circuit 112 as the T1 signal (AC lightsignal). The signal receiving circuit 112 converts the T1 signal as theAC light signal into T2 signal as the DC voltage signal. The voltagelevel of the T2 signal is directly proportional to the correctionvoltage.

The current phase detection circuit 144 adjusts an S2 signal (AC voltagesignal) indicating a current phase by using the T2 signal (DC voltagesignal) indicating the correction voltage and outputs the S1 signal (ACvoltage signal) as a correction current phase. When the T2 signal iszero, that is, when the load voltage V5 assumes a desired value, the S2signal directly becomes the S1 signal. The phase comparison circuit 150detects the phase difference between the voltage phase and current phaseof the AC power based on the S0 and S1 signals and outputs the phasedifference indicating voltage SC. The VCO 202 adjusts the drivefrequency fo based on the phase difference indicating voltage SC. Morespecifically, the VCO 202 changes the pulse width of the AC voltage Voto thereby change the drive frequency fo.

Also, when the correction voltage, i.e., T2 signal is not zero, thephase comparison circuit 150 detects the phase difference between thevoltage phase and current phase of the AC power based on the S0 and S1signals and outputs the phase difference indicating voltage SC. However,the S1 signal at this time is a signal obtained by correcting the S2signal in accordance with the T2 signal, so that the S1 signal does notindicate the actual current phase. The adjustment logic based on thecorrection voltage will be described later using FIG. 12.

A detection coil LSS is provided at the feeding coil circuit 120. Thedetection coil LSS is a coil wound around a core 154 (toroidal core)having a penetration hole NS times. The core 154 is formed of a knownmaterial such as ferrite, silicon steel, or permalloy. The number ofwindings NS of the detection coil LSS in the present embodiment is 100.

A part of the current path of the feeding coil circuit 120 penetratesthe penetration hole of the core 154. This means that the number ofwindings NP of the feeding coil circuit 120 with respect to the core 154is one. With the above configuration, the detection coil LSS and feedingcoil L2 constitute a coupling transformer. An AC magnetic fieldgenerated by the AC current I2 of the feeding coil L2 causes inductivecurrent ISS having the same phase as that of the current I2 to flow inthe detection coil LSS. The magnitude of the inductive current ISS isrepresented by I2·(NP/NS) according to the law of equal ampere-turn.

A resistor R4 is connected to both ends of the detection coil LSS. Oneend B of the resistor R4 is grounded, and the other end A thereof isconnected to the current phase detection circuit 144 through acomparator 142.

Potential VSS is digitized by the comparator 142 to be an S2 signal. Thecomparator 142 outputs a saturated voltage of 3.0 (V) when the potentialVSS exceeds a predetermined threshold, e.g., 0.1 (V). Thus, thepotential VSS is converted into the S2 signal of a digital waveform bythe comparator 142. The current I2 and inductive current ISS have thesame phase, and inductive current ISS and potential VSS (S2 signal) havethe same phase. Further, the AC current Is flowing in the powertransmission control circuit 200 have the same phase as that of thecurrent I2. Therefore, by observing the waveform of the S2 signal, thecurrent phase of the AC current Is can be measured.

FIG. 3 is a graph illustrating a relationship between load current I5and load voltage V5. The horizontal axis represents the magnitude of theload current I5 (DC) flowing in the load LD, and the vertical axisrepresents the load voltage V5. A non-adjustment characteristic 134represents a current-voltage characteristic obtained in the case whereadjustment based on the correction voltage is not performed. In the caseof the non-adjustment characteristic 134, when the load LD increases,the load current I5 decreases while the load voltage V5 increases. Onthe other hand, when the load LD decreases, the load current I5increases while the load voltage V5 decreases. As described above, whenthe load LD changes, the load voltage V5 changes even when constantpower is fed.

The wireless power transmission system 100 in the first embodimentachieves the current-voltage characteristic represented by an adjustmentcharacteristic 136. To be specific, the S1 signal is adjusted based onthe correction voltage to change the power transmission efficiency,whereby the load voltage V5 is stabilized.

FIG. 4 is a graph illustrating a relationship between inter-coildistance d and load voltage V5. The horizontal axis represents theinter-coil distance d between the feeding coil L2 and receiving coli L3,and the vertical axis represents the load voltage V5. A non-adjustmentcharacteristic 146 represents a voltage-distance characteristic obtainedin the case where adjustment based on the correction voltage is notperformed. As describe above, the resonance frequency fr1 changesdepending on the inter-coil distance d. When the resonance frequency fr1changes to cause the difference between the drive frequency fo andresonance frequency fr1 to change, the power transmission efficiencychanges. Even when the drive frequency fo is made to track the resonancefrequency fr1, the load voltage V5 changes to a certain degree dependingon the inter-coil distance d.

The wireless power transmission system 100 in the first embodimentachieves the voltage-distance characteristic represented by anadjustment characteristic 148. That is, the S1 signal is adjusted basedon the correction voltage to change the power transmission efficiency,whereby the load voltage V5 is stabilized.

FIG. 5 is a graph illustrating a relationship between the impedance Z ofthe feeding coil circuit 120 and drive frequency fo. The vertical axisrepresents the impedance Z of the feeding coil circuit 120 (a circuit inwhich the capacitor C2 and the feeding coil L2 are connected in series).The horizontal axis represents the drive frequency fo. The impedance Zis a minimum value Zmin at the resonance state. Although Zmin=0 at theresonance state is ideal, Zmin does not become zero in general sincesome resistance components are included in the feeding coil circuit 120.

When the drive frequency fo and resonance frequency fr1 coincide witheach other, the impedance Z becomes minimum and the capacitor C2 and thefeeding coil L2 are in a resonance state. When the drive frequency foand resonance frequency fr1 deviate from each other, one of thecapacitive reactance and inductive reactance prevails the other, so thatthe impedance Z is also increased.

The impedance Z increases as the deviation from the drive frequency foand resonance frequency fr1 advances, with the result that the powertransmission efficiency is degraded. That is, it is possible to changethe power transmission efficiency by changing the difference between thedrive frequency fo and resonance frequency fr1.

FIG. 6 is a graph illustrating a relationship between the output powerefficiency and drive frequency fo. The output power efficiency is aratio of power actually fed from the feeding coil L2 relative to themaximum output value. When the drive frequency fo coincides with theresonance frequency fr1, a difference between the current phase andvoltage phase becomes zero and therefore the power transmissionefficiency becomes maximum, with the result that output power efficiencyof 100(%) can be obtained. In the wireless power transmission system 100of the first embodiment, the drive frequency fo is adjusted in a rangeof fo1 to fo2 which is lower than the resonance frequency fr1.

FIG. 7 is a circuit diagram of the signal transmission circuit 122 inthe first embodiment. The signal transmission circuit 122 includes aninfrared ray LED (Light Emitting Diode) 158, a transistor Q3, and a VFconverter 156. The transistor Q3 is an emitter-grounded bipolartransistor, and the base and emitter thereof are connected through aresistor R6. One end of the infrared ray LED 158 is connected to thepower supply VCC through the resistor R5, and the other end thereof isconnected to the collector of the transistor Q3. The VF converter 156 isalso connected to the base of the transistor Q3.

The measurement circuit 126 transmits the T0 signal (DC voltage signal)indicating the correction voltage to the VF converter 156. The VFconverter 156 generates a T3 signal (AC voltage signal) which is a pulsesignal having a duty ratio of 50%. A signal frequency fs1 of the T3signal changes depending on the T0 signal (correction voltage). FIG. 8is a graph illustrating a relationship between the signal frequency fs1in the VF converter 156 and T0 signal.

The T3 signal (AC voltage signal) is changed into the T1 signal (AClight signal) by the infrared ray LED 158. The infrared ray LED 158transmits the T1 signal (AC light signal) to the signal receivingcircuit 112. The T1 signal is a light signal (infrared ray signal) thatblinks at the signal frequency fs1. The wavelength of the T1 signal isabout 940 nm. The T1 signal travels up to several meters, so that thereoccurs no problem even if the inter-coil distance d is large. Further,the infrared ray is hardly subject to the magnetic field generated bythe feeding coil L2, an advantage that the T1 signal and feeding powerhardly interact with each other can be obtained.

FIG. 9 is a circuit diagram of the signal receiving circuit 112 in thefirst embodiment. The signal receiving circuit 112 includes a photodiode160, a voltage conversion section 164, and an FV converter 138. Thevoltage conversion section 164 includes a comparator 162 and a resistorR7.

The photodiode 160 receives the T1 signal (AC light signal) of thesignal frequency fs1. The T1 signal (AC light signal) is converted intoa T4 signal (AC voltage signal) by the voltage conversion section 164.In the voltage conversion section 164, the resistor R7 is adjusted inthe way output 1 (mV) per 1 lux. The brightness of the T1 signal at thereception time is about 0 to 2000 (lux) and, accordingly, the voltagelevel of the T4 signal is 0 to 2.0 (V). The T4 signal is a pulse-likevoltage signal changing at the signal frequency fs1. The duty ratio ofthe T4 signal is 50%. The T4 signal (AC voltage signal) is convertedinto the fixed T2 value (DC voltage signal) by the FV converter 138.FIG. 10 is a graph illustrating a relationship between the signalfrequency fs1 in the FV converter 138 and T2 signal. The higher thesignal frequency fs1, the higher the voltage level of the T2 signal isset.

FIG. 11 is a circuit diagram of the current phase detection circuit 144in the first embodiment. The current phase detection circuit 144includes a comparator 166 and a current waveform shaping circuit 168.The potential VSS is shaped into the S2 signal of a digital waveform bythe comparator 142 and input to the current waveform shaping circuit168. The current waveform shaping circuit 168 shapes the S2 signal of adigital waveform (rectangular waveform) into an S3 signal of a saw-toothwaveform. In the current waveform shaping circuit 168, a resistor R8 isinserted in the path of the S2 signal, and a diode D1 is connected inparallel to the resistor R8. The transmission path of the S2 signal isgrounded through a capacitor C7.

The S3 signal (AC voltage signal) is input to the positive terminal ofthe comparator 166, and T2 signal (DC voltage signal) output from thesignal receiving circuit 112 is input to the negative terminal of thecomparator 166. The S3 signal is a signal indicating a current phase,and T2 signal is a DC voltage signal indicating the correction voltage.

The comparator 166 outputs a high-level S1 signal when the level of theS3 signal is higher than that of the T2 signal while it outputs alow-level S1 signal in the rest of the time. The actual outputwavelength will be described in detail later using FIG. 12 andsubsequent drawings.

FIG. 12 is a time chart illustrating a relationship between the S0signal and S1 signal. Time period from time t0 to time t1 (hereinafter,referred to as “first period”) is a time period during which theswitching transistor Q1 is ON while the switching transistor Q2 is OFF.Time period from time t1 to time t2 (hereinafter, referred to as “secondperiod”) is a time period during which the switching transistor Q1 isOFF while the switching transistor Q2 is ON. Time period from time t2 totime t3 (hereinafter, referred to as “third period”) is a time periodduring which the switching transistor Q1 is ON while the switchingtransistor Q2 is OFF. Time period from time t3 to time t4 (hereinafter,referred to as “fourth period”) is a time period during which theswitching transistor Q1 is OFF while the switching transistor Q2 is ON.

At time t0, the AC voltage Vo (S0 signal) changes from the minimum valueof 0.0 (V) to the maximum value of 3.0 (V). At time t1 at which thefirst time period is ended, the AC voltage Vo (S0 signal) changes fromthe maximum value of 3.0 (V) to the minimum value of 0.0 (V).Hereinafter, a timing (represented by, e.g., time t0) at which the S0signal rises is referred to as “voltage phase value”.

In the case where the drive frequency fo is lower than the resonancefrequency fr1, a capacitive reactance component appears in the impedanceZ of the feeding coil circuit 120 (LC resonance circuit), and thecurrent phase of the current Is advances with respect to the voltagephase. Thus, the S2 signal indicating a current phase rises at time t6which is earlier than time t0. Hereinafter, a timing (represented by,e.g., time t6) at which the S2 signal rises is referred to as “currentphase value”. In the example of FIG. 12, a value obtained by t0-t6represents the phase difference. Here, t0-t6>0 is established, so thatthe current phase advances with respect to the voltage phase.

When the S2 signal rises at time t6, the level of the S3 signal startsincreasing. At time t8 at which the level of the S2 signal becomes zero,the level of the S3 signal also abruptly decreases from the maximumvalue of 3.0 (V) to 0.0 (V).

The T2 signal is a DC voltage signal whose level changes depending onthe magnitude of the correction voltage. In FIG. 12, the correctionvoltage is detected, that is, the load voltage V5 deviates from adesired value.

The S3 signal and T2 signal are input to the positive terminal andnegative terminals of the comparator 166, respectively, and the S1signal is output from the comparator 166. During the period during whichthe level of the S3 signal is higher than that of T2 signal (S3>T2), thelevel of the S1 signal is higher than 0 (S1>0), while in the rest oftime, the level of the S1 signal is 0 (S1=0). In FIG. 12, the level ofthe S3 signal is higher than that of the T2 (S3>T2) at time t7(hereinafter, such a timing is referred to also as “corrected currentphase value”) which is later than time t0. The voltage level of the T2signal serves as a “reference value” for determining the correctedcurrent phase value.

The phase comparison circuit 150 compares rising edge time t0 of the S0signal and rising edge time t7 of the S1 signal to calculate the phasedifference td. Although the actual phase difference is t0-t6 (>0), thephase difference recognized by the phase comparison circuit 150 isobtained by t0-t7 (<0). The phase comparison circuit 150 outputs thephase difference indicating voltage SC corresponding to a value obtainedby t0-t7. The VCO 202 determines that the current phase delays withrespect to the voltage phase, that is, the drive frequency fo is higherthan the resonance frequency fr1 and tries to eliminate the phasedifference by reducing the drive frequency fo. As a result, feedbackcontrol is executed such that the power transmission efficiency isdegraded, the load voltage V5 is reduced, and the correction voltage iseliminated.

FIG. 13 is a circuit diagram illustrating a modification example of thewireless power receiver 118 in the first embodiment. Although the DCcurrent I5 is fed to the load LD in FIG. 2, the AC current I4 maydirectly be fed to the load LD in the modification example. In thiscase, the rectification circuit 124 and measurement circuit 126 areconnected to a part of the load coil L4 so as to allow the T0 signal tobe output.

FIG. 14 is a system configuration view of a wireless power transmissionsystem 100 which is a modification of the first embodiment. In thewireless power transmission system 100 of the modification, the powertransmission control circuit 200 directly drives the feeding coilcircuit 120 without intervention of the exciting circuit 110. Componentsdesignated by the same reference numerals as those of FIG. 2 have thesame or corresponding functions as those in FIG. 2.

The feeding coil circuit 120 in the modification is a circuit in whichthe transformer TS2 secondary coil Li is connected in series to thefeeding coil L2 and capacitor C2. The transformer TS2 secondary coil Liconstitutes a coupling transformer TS2 together with the transformer TS2primary coil Lb and receives AC power from the power transmissioncontrol circuit 200 by electromagnetic induction. Thus, the AC power maybe directly fed from the power transmission control circuit 200 to thefeeding coil circuit 120 without intervention of the exciting circuit110.

Although the power transmission control circuit 200 is a half-bridgetype circuit, it may be constructed as a push-pull type circuit orfull-bridge type circuit. The S3 signal generated by the currentwaveform shaping circuit 168 may be an AC signal having not only asaw-tooth waveform but also a triangle wave or a sine wave in which avoltage value is gradually increased or decreased within a predeterminedtime period. Although the current phase is set as an adjustment targetin the first embodiment, the voltage phase may be adjusted based on theT0 signal. Further, the feedback control may be effected based not onlyon the output voltage but on the current or power.

Although the signal frequency fs1 of the T1 signal (AC light signal)indicates the correction voltage in the first embodiment, the correctionvoltage may be represented by the amplitude or duty ratio of the T1signal. Alternatively, numerical information indicating the correctionvoltage may be transmitted as a digital signal. The T1 signal is notlimited to a light signal such as an infrared ray but may be a radiosignal. At any rate, it is only necessary for the T1 signal to have afrequency band sufficiently away from the frequency band of the drivefrequency fo or resonance frequency fr1. The infrared ray LED 158 andphotodiode 160 are comparatively low in price, so that the light signalis adopted in the first embodiment.

Second Embodiment

FIG. 15 is a system configuration view of a wireless power transmissionsystem 1100 according to the second embodiment. The wireless powertransmission system 1100 includes a wireless power feeder 1116 and awireless power receiver 1118. The wireless power feeder 1116 includes,as basic components, a power transmission control circuit 1200, anexciting circuit 1110, a feeding coil circuit 1120, a phase detectioncircuit 1114, and a signal receiving circuit 1112. The wireless powerreceiver 1118 includes a receiving coil circuit 1130, a loading circuit1140, a control signal generation circuit 1170, a reference signalgeneration circuit 1172, and a signal transmission circuit 1122.

A distance (hereinafter, referred to as “inter-coil distance”) of about0.2 m to 1.0 m is provided between a feeding coil L2 of the feeding coilcircuit 1120 and a receiving coil L3 of the receiving coil circuit 1130.The wireless power transmission system 1100 mainly aims to feed powerfrom the feeding coil L2 to receiving coil L3 by wireless. In the secondembodiment, a description will be made assuming that resonance frequencyfr1 is 100 kHz. The wireless power transmission system of the secondembodiment may be made to operate in a high-frequency band like ISM(Industry-Science-Medical) frequency band. A low frequency band isadvantageous over a high frequency band in reduction of cost of aswitching transistor (to be described later) and reduction of switchingloss. In addition, the low frequency band is less constrained by RadioAct.

The exciting circuit 1110 is a circuit in which an exciting coil L1 anda transformer TS2 secondary coil Li are connected in series. Thetransformer TS2 secondary coil Li constitutes a coupling transformer TS2together with a transformer TS2 primary coil Lb and receives AC powerfrom the power transmission control circuit 1200 by electromagneticinduction. The number of windings of the exciting coil L1 is 1, diameterof a conductive wire is 5 mm, and shape of the exciting coil L1 itselfis a square of 210 mm×210 mm. In FIG. 15, the exciting coil L1 isrepresented by a circle for clarification. Other coils are alsorepresented by circles for the same reason. All the coils illustrated inFIG. 15 are made of copper. Current I1 flowing in the exciting circuit1110 is AC.

The feeding coil circuit 1120 is a circuit in which a feeding coil L2and a capacitor C2 are connected in series. The exciting coil L1 andfeeding coil L2 face each other. The distance between the exciting coilL1 and feeding coil L2 is as comparatively small as 10 mm or less. Thus,the exciting coil L1 and feeding coil L2 are electromagneticallystrongly coupled to each other. The number of windings of the feedingcoil L2 is 7, diameter of a conductive wire is 5 mm, and shape of thefeeding coil L2 itself is a square of 280 mm×280 mm. When the AC currentI1 is made to flow in the exciting coil L1, an electromotive forceoccurs in the feeding coil L2 according to the principle ofelectromagnetic induction to cause AC current I2 to flow in the feedingcoil circuit 1120. The AC current I2 is considerably larger than the ACcurrent I1. The values of the feeding coil L2 and capacitor C2 are setsuch that the resonance frequency fr1 of the feeding coil circuit 1120is 100 kHz.

The receiving coil circuit 1130 is a circuit in which a receiving coilL3 and a capacitor C3 are connected in series. The feeding coil L2 andreceiving coil L3 face each other. The number of windings of thereceiving coil L3 is 7, diameter of a conductive wire is 5 mm, and shapeof the receiving coil L3 itself is a square of 280 mm×280 mm. The valuesof the receiving coil L3 and capacitor C3 are set such that theresonance frequency fr1 of the receiving coil circuit 1130 is also 100kHz. Thus, the feeding coil L2 and receiving coil L3 need not have thesame shape. When the feeding coil L2 generates a magnetic field at theresonance frequency fr1=100 kHz, the feeding coil L2 and receiving coilL3 magnetically resonate, causing large current I3 to flow in thereceiving coil circuit 1130.

The loading circuit 1140 is a circuit in which a loading coil L4 isconnected to a load LD through a rectification circuit 1124 and ameasurement circuit 1126. The receiving coil L3 and loading coil L4 faceeach other. The distance between the receiving coil L3 and loading coilL4 is as comparatively small as about 10 mm or less. Thus, the receivingcoil L3 and loading coil L4 are electromagnetically strongly coupled toeach other. The number of windings of the loading coil L4 is 1, diameterof the wire of the loading coil L4 is 5 mm, and shape of the loadingcoil L4 itself is a square of 300 mm×300 mm. When the current I3 is madeto flow in the receiving coil L3, an electromotive force occurs in theloading circuit 1140 to cause AC current I4 to flow in the loadingcircuit 1140. The AC current I4 is rectified into DC current by therectification circuit 1124. Although part of the DC current flows in themeasurement circuit 1126, most of the DC current flows in the load LD asDC current I5. The rectification circuit 1124 is a general circuitconstituted by a bridge circuit 1128 and a capacitor C5. The details ofthe measurement circuit 1126 will be described later.

The AC power fed from the feeding coil L2 of the wireless power feeder1116 is received by the receiving coil L3 of the wireless power receiver1118 and then extracted from the load LD as DC power. Voltage applied tothe load LD is referred to as “load voltage V5”.

If the load LD is connected in series to the receiving coil circuit1130, the Q-value of the receiving coil circuit 1130 is degraded.Therefore, the receiving coil circuit 1130 for power reception andloading circuit 1140 for power extraction are separated from each other.In order to enhance the power transmission efficiency, the center linesof the feeding coil L2, receiving coil L3, and loading coil L4 arepreferably made to coincide with one another.

The measurement circuit 1126 includes resistors R1 and R2, a controlpower supply Vs, and a comparator 1132. The load voltage V5 is dividedby the resistors R1 and R2. Voltage applied to both ends of theresistors R2 is referred to as “output voltage”. The potential at aconnecting point F between the resistors R1 and R2 is input to thenegative terminal of the comparator 1132 as “measurement potential”. Acontrol power supply Vs is connected to the positive terminal of thecomparator 1132. Input voltage at the positive terminal of thecomparator 1132 generated by the control power supply Vs is referred toas “reference potential”.

The comparator 1132 amplifies a difference (hereinafter, referred to as“correction voltage”) between the measurement potential and referencepotential and outputs the amplified value as a T0 signal. The T0 signalis a DC voltage signal and indicates the magnitude of the correctionvoltage. In other words, the T0 signal is a signal indicating a changein the load voltage V5. Although the details will be described later, inthe wireless power transmission system 1100 of the second embodiment,the feeding power is controlled so as to make the correction voltage bezero to thereby stabilize the output voltage (load voltage V5). In thesecond embodiment, the reference potential is set to 2.5 (V). Further,the resistors R1 and R2 are set such that the measurement potential is2.5 (V) and correction voltage is 0 (V) when the load voltage V5 is 24(V). The control voltage Vs is a variable DC voltage supply and canarbitrarily be adjusted.

The control signal generation circuit 1170 is a circuit that generatesan AC voltage signal of a control frequency fc as a T4 signal. Thecontrol frequency fc in the second embodiment is 1.0 kHz. The details ofthe control signal generation circuit 1170 will be described later usingFIG. 20. The comparator 1174 compares the T0 signal and T4 signal andgenerates a high-level T2 signal (enable signal: AC voltage signal) whenthe level of the T4 signal is higher than that of the T0 signal (T4>T0).Although details will be described later, the duty ratio of the T2signal changes depending on the correction voltage. A relationship amongthe T0, T2 and T4 signals will be described later using FIG. 21.

The reference signal generation circuit 1172 generates an AC voltagesignal of a reference frequency fs as a T3 signal. The referencefrequency fs in the second embodiment is 38 kHz. The signal transmissioncircuit 1122 generates a T1 signal as an AC light signal based on the T2and T3 signals. The T1 signal is an “output signal” indicating themagnitude of the output on the power receiving side and is received bythe signal receiving circuit 1112 of the wireless power feeder 1116.Based on the T1 signal, the power feeding side can recognize themagnitude of the correction voltage, in other words, a variation of theload voltage V5. Circuit configurations and processing of the signaltransmission circuit 1122 and reference signal generation circuit 1172will be described later using FIGS. 22 and 23.

A configuration of the power transmission control circuit 1200 will bedescribed. A VCO (Voltage Controlled Oscillator) 1202 is connected tothe primary side of the gate-drive transformer TS1. The VCO 1202functions as an “oscillator” that generates AC voltage Vo at the drivefrequency fo. Although the waveform of the AC voltage Vo may be a sinewave, it is assumed here that the voltage waveform is a rectangular wave(digital wave). The AC voltage Vo causes current to flow in atransformer TS1 primary coil Lh alternately in both positive andnegative directions. A transformer TS1 primary coil Lh, a transformerTS1 secondary coil Lf, and a transformer TS1 secondary coil Lgconstitute a gate-drive coupling transformer TS1. Electromagneticinduction causes current to flow also in the transformer TS1 secondarycoil Lf and transformer TS1 secondary coil Lg alternately in bothpositive and negative directions.

As the VCO 1202 in the second embodiment, a built-in unit (productserial number MC14046B) manufactured by Motorola, Inc is used. The VCO1202 also has a function of dynamically changing the drive frequency fobased on phase difference indicating voltage SC fed from the phasecomparison circuit 1150 (described later in detail).

The following description will be made assuming that the minimum valuefo1 of the drive frequency fo is 90 kHz, and the maximum value fo2thereof is 99 kHz. The appropriate range of the phase differenceindicating voltage SC is 1.0 (V) to 4.0 (V). The phase differenceindicating voltage SC and drive frequency fo are directly proportionalto each other. That is, when the phase difference indicating voltage SCis 1.0 (V), the drive frequency fo (=fo1) is 90 kHz, and when the phasedifference indicating voltage SC is 4.0 (V), the drive frequency fo(=fo2) is 99 kHz.

Capacitors CA and CB charged by a DC power supply Vdd each serve as apower supply for the power transmission control circuit 1200. Thecapacitor CA is provided between points C and E of FIG. 15, andcapacitor CB is provided between points E and D. Assuming that thevoltage (voltage between points C and E) of the capacitor CA is VA,voltage (voltage between points E and D) of the capacitor CB is VB,VA+VB (voltage between points C and D) represents input voltage. Thatis, the capacitors CA and CB each function as a DC voltage supply.

One end of the transformer TS1 secondary coil Lf is connected to thegate of a switching transistor Q1, and the other end of the transformerTS1 secondary coil Lf is connected to the source of a switchingtransistor Q1. One end of the transformer TS1 secondary coil Lg isconnected to the gate of a switching transistor Q2, and the other end ofthe transformer TS1 secondary coil Lg is connected to the source of aswitching transistor Q2. When VCO 1202 generates AC voltage Vo at drivefrequency fo, voltage Vx (Vx>0) is alternately applied, at drivefrequency fo, to the gates of the switching transistors Q1 and Q2. As aresult, the switching transistors Q1 and Q2 are alternately turnedon/off at the drive frequency fo. The switching transistors Q1 and Q2are enhancement type MOSFET (Metal Oxide Semiconductor Field effecttransistor) having the same characteristics but may be other transistorssuch as a bipolar transistor. Further, other switches such as a relayswitch may be used in place of the transistor.

The drain of the switching transistor Q1 is connected to the positiveelectrode of the capacitor CA. The negative electrode of the capacitorCA is connected to the source of the switching transistor Q1 through thetransformer TS2 primary coil Lb. The source of the switching transistorQ2 is connected to the negative electrode of the capacitor CB. Thepositive electrode of the capacitor CB is connected to the drain of theswitching transistor Q2 through the transformer TS2 primary coil Lb.

Voltage between the source and drain of the switching transistor Q1 isreferred to as source-drain voltage VDS1, and voltage between the sourceand drain of the switching transistor Q2 is referred to as source-drainvoltage VDS2. Current flowing between the source and drain of theswitching transistor Q1 is referred to as source-drain current IDS1, andcurrent flowing between the source and drain of the switching transistorQ2 is referred to as source-drain current IDS2. The directions of arrowsin the diagram indicate the positive directions, and directions oppositeto the directions of the arrows indicate the negative directions.

When the switching transistor Q1 is turned conductive (ON), theswitching transistor Q2 is turned non-conductive (OFF). A main currentpath (hereinafter, referred to as “first current path”) at this timeextends from the positive electrode of the capacitor CA, passes throughthe point C, switching transistor Q1, transformer TS2 primary coil Lb,and point E in this order, and returns to the negative electrode of thecapacitor CA. The switching transistor Q1 functions as a switch forcontrolling conduction/non-conduction of the first current path.

When the switching transistor Q2 is turned conductive (ON), theswitching transistor Q1 is turned non-conductive (OFF). A main currentpath (hereinafter, referred to as “second current path”) at this timeextends from the positive electrode of the capacitor CB, passes throughthe point E, transformer TS2 primary coil Lb, switching transistor Q2,and point D in this order, and returns to the negative electrode of thecapacitor CB. The switching transistor Q2 functions as a switch forcontrolling conduction/non-conduction of the second current path.

Current flowing in the transformer TS2 primary coil Lb in the powertransmission control circuit 1200 is referred to as “current IS”. Thecurrent IS is AC current, and the current flow in a first current pathis defined as the positive direction and current flow in a secondcurrent path is defined as the negative direction.

When the VCO 1202 feeds the AC voltage Vo at the drive frequency fo, thefirst and second current paths are switched at the drive frequency fo.Since the AC current Is of the drive frequency fo flows in thetransformer TS2 primary coil Lb, the AC current I1 flows in the excitingcircuit 1110 at the drive frequency fo, and the AC current I2 of thedrive frequency fo flows in the feeding circuit 1120. The closer thevalue of the drive frequency fo is to the resonance frequency fr1, thehigher the power transmission efficiency becomes. When the drivefrequency fo is equal to the resonance frequency fr1, the feeding coilL2 of the feeding coil circuit 1120 and capacitor C2 are in a resonancestate. The receiving coil circuit 1130 is also a resonance circuit ofthe resonance frequency fr1, so that the feeding coil L2 and receivingcoil L3 magnetically resonate. At this time, the maximum transmissionefficiency can be obtained.

In the case of the second embodiment, however, the resonance frequencyfr1 is not included in the operating range of the drive frequency fo, sothat the power transmission efficiency does not reach the maximum value.This is because priority is given to the stability of the load voltageV5 over the maximization of the power transmission efficiency. A changein the load voltage V5 can be detected from the correction voltage, sothat the wireless power feeder 1116 automatically adjusts the drivefrequency fo so as to make the correction voltage be zero. The detailswill be described later.

The resonance frequency fr1 slightly changes depending on use conditionor use environment of the feeding coil circuit 1120 or receiving coilcircuit 1130. Further, in the case where the feeding coil circuit 1120or receiving coil circuit 1130 is replaced with new one, the resonancefrequency fr1 changes. Alternatively, there may be case where theresonance frequency fr1 needs to be changed aggressively by setting theelectrostatic capacitance of the capacitor C2 or capacitor C3 variable.Further, according to the experiment made by the present inventor, ithas been found that the resonance frequency fr1 starts falling when thedistance between the feeding coil L2 and receiving coil L3 is madesmaller to some extent. When the difference between the resonancefrequency fr1 and drive frequency fo changes, the power transmissionefficiency changes. When the power transmission efficiency changes, theload voltage V5 also changes. Therefore, in order to stabilize the loadvoltage V5, it is necessary to keep the difference between the resonancefrequency fr1 and the drive frequency fo constant even if the resonancefrequency fr1 changes.

A detection coil LSS is provided at the feeding coil circuit 1120. Thedetection coil LSS is a coil wound around a core 1154 (toroidal core)having a penetration hole NS times. The core 1154 is formed of a knownmaterial such as ferrite, silicon steel, or permalloy. The number ofwindings NS of the detection coil LSS in the present embodiment is 100.

A part of the current path of the feeding coil circuit 1120 penetratesthe penetration hole of the core 1154. This means that the number ofwindings NP of the feeding coil circuit 1120 with respect to the core1154 is one. With the above configuration, the detection coil LSS andfeeding coil L2 constitute a coupling transformer. An AC magnetic fieldgenerated by the AC current I2 of the feeding coil L2 causes inductivecurrent ISS having the same phase as that of the current I2 to flow inthe detection coil LSS. The magnitude of the inductive current ISS isrepresented by I2·(NP/NS) according to the law of equal ampere-turn.

A resistor R4 is connected to both ends of the detection coil LSS. Oneend B of the resistor R4 is grounded, and the other end A thereof isconnected to the current phase detection circuit 1144 through acomparator 1142.

Potential VSS is digitized by the comparator 1142 to be an S2 signal.The comparator 1142 outputs a saturated voltage of 3.0 (V) when thepotential VSS exceeds a predetermined threshold, e.g., 0.1 (V). Thus,the potential VSS is converted into the S2 signal of a digital waveformby the comparator 1142. The current I2 and inductive current ISS havethe same phase, and inductive current ISS and potential VSS (S2 signal)have the same phase. Further, the AC current Is flowing in the powertransmission control circuit 1200 have the same phase as that of thecurrent I2. Therefore, by observing the waveform of the S0 signal, thecurrent phase of the AC current Is can be measured.

When the resonance frequency fr1 and drive frequency fo coincide witheach other, the current phase and voltage phase also coincide with eachother. A deviation between the resonance frequency fr1 and drivefrequency fo can be measured from the phase difference between thecurrent phase and voltage phase. The wireless power transmission system1100 in the present embodiment measures the deviation between theresonance frequency fr1 and drive frequency fo based on the phasedifference to thereby make the drive frequency fo automatically track achange of the resonance frequency fr1.

The phase detection circuit 1114 includes a current phase detectioncircuit 1144, a phase comparison circuit 1150, and a low-pass filter1152. The low-pass filter 1152 is a known circuit and inserted so as tocut a high-frequency component of the phase difference indicatingvoltage SC. As the phase comparison circuit 1150 in the secondembodiment, a built-in unit (Phase Comparator) (product serial numberMC14046B) manufactured by Motorola is used, as in the case of the VCO1202. Thus, the phase comparison circuit 1150 and VCO 1202 can beimplemented in one chip.

The current phase detection circuit 1144 generates an S1 signal as asignal indicating a current phase. The S1 signal is input to the phasecomparison circuit 1150. The AC voltage Vo generated by the VCO 1202 isinput to the phase comparison circuit 1150 as an S0 signal indicating avoltage phase. The phase comparison circuit 1150 detects a deviation(phase difference) between the current phase and voltage phase from theS1 and S0 signals and generates the phase difference indicating voltageSC indicating the magnitude of the phase difference. Detecting the phasedifference allows detection of the magnitude of the deviation betweenthe resonance frequency fr1 and drive frequency fo. It is possible tokeep the phase difference between the drive frequency fo and theresonance frequency fr1 constant by controlling the drive frequency foaccording to the phase difference indicating voltage SC.

For example, when the drive frequency fo and resonance frequency fr1deviate from each other, the phase difference is accordingly increased,so that the phase comparison circuit 1150 generates the phase differenceindicating voltage SC so as to reduce the phase difference. Thus, evenif the resonance frequency fr1 changes, it is possible to keep the powertransmission efficiency constant to thereby stabilize the load voltageV5. A circuit configuration of the current phase detection circuit 1144and the signal receiving circuit 1112 will be described later using FIG.24, and relationship between the S1 and S3 (S2) signals will bedescribed later using FIG. 25.

The S0 signal may be obtained by connecting resistors to both ends ofthe transformer TS1 primary coil Lh in parallel to divide the AC voltageVo. Even in the case where the AC voltage Vo generated by the VCO 1202is large, the AC voltage can be reduced to a manageable level by thevoltage division. The voltage phase may be measured from thesource-drain voltages VDS1 and VDS2 or source-gate voltages VGS1 andVGS2.

Even though the resonance frequency fr1 is constant, the load voltage V5may be changed in some cases. For example, in the case where the load LDis a variable resistor or in the case where the load LD is replaced withnew one, the load voltage V5 changes. In the first embodiment, a changein the load voltage V5 is detected as the correction voltage, and thedrive frequency fo is automatically adjusted so as to make thecorrection voltage be zero, whereby the load voltage V5 is stabilized.

The correction voltage is transmitted from the signal transmissioncircuit 1122 to signal receiving circuit 1112 as the T1 signal (AC lightsignal). The signal receiving circuit 1112 converts the T1 signal as theAC light signal into T5 signal as the DC voltage signal. The voltagelevel of the T5 signal is directly proportional to the load voltage V5.

The current phase detection circuit 1144 adjusts an S2 signal (ACvoltage signal) indicating a current phase by using the T5 signal (DCvoltage signal) indicating the correction voltage and outputs the S1signal (AC voltage signal) as a correction current phase. When the loadvoltage V5 assumes a desired value 24 (V), the S2 signal directlybecomes the S1 signal. The phase comparison circuit 1150 detects thephase difference between the voltage phase and current phase of the ACpower based on the S1 and S0 signals and outputs the phase differenceindicating voltage SC. The VCO 1202 adjusts the drive frequency fo basedon the phase difference indicating voltage SC. More specifically, theVCO 1202 changes the pulse width of the AC voltage Vo to thereby changethe drive frequency fo.

Also, when the correction voltage is not zero, that is, when S1 signalis adjusted by T5 signal, the phase comparison circuit 1150 detects thephase difference between the voltage phase and current phase of the ACpower based on the S1 and S0 signals and outputs the phase differenceindicating voltage SC. However, the S1 signal at this time is a signalobtained by correcting the S2 signal in accordance with the T5 signal,so that the S1 signal does not indicate the actual current phase. Theadjustment logic based on the correction voltage will be described laterusing FIG. 26.

FIG. 16 is a graph illustrating a relationship between load current I5and load voltage V5. The horizontal axis represents the magnitude of theload current I5 (DC) flowing in the load LD, and the vertical axisrepresents the load voltage V5. A non-adjustment characteristic 1134represents a current-voltage characteristic obtained in the case whereadjustment based on the correction voltage is not performed. In the caseof the non-adjustment characteristic 1134, when the load LD increases,the load current I5 decreases while the load voltage V5 increases. Onthe other hand, when the load LD decreases, the load current I5increases while the load voltage V5 decreases. As described above, whenthe load LD changes, the load voltage V5 changes even when constantpower is fed.

The wireless power transmission system 1100 in the second embodimentachieves the current-voltage characteristic represented by an adjustmentcharacteristic 1136. To be specific, the S1 signal is adjusted based onthe correction voltage to change the power transmission efficiency,whereby the load voltage V5 is stabilized.

FIG. 17 is a graph illustrating a relationship between inter-coildistance d and load voltage V5. The horizontal axis represents theinter-coil distance d between the feeding coil L2 and receiving coil L3,and the vertical axis represents the load voltage V5. A non-adjustmentcharacteristic 1146 represents a voltage-distance characteristicobtained in the case where adjustment based on the correction voltage isnot performed. As describe above, the resonance frequency fr1 changesdepending on the inter-coil distance d. When the resonance frequency fr1changes to cause the difference between the drive frequency fo andresonance frequency fr1 to change, the power transmission efficiencychanges. Even when the drive frequency fo is made to track the resonancefrequency fr1, the load voltage V5 changes to a certain degree dependingon the inter-coil distance d.

The wireless power transmission system 1100 in the second embodimentachieves the voltage-distance characteristic represented by anadjustment characteristic 1148. That is, the S1 signal is adjusted basedon the correction voltage to change the power transmission efficiency,whereby the load voltage V5 is stabilized.

FIG. 18 is a graph illustrating a relationship between the impedance Zof the feeding coil circuit 1120 and drive frequency fo. The verticalaxis represents the impedance Z of the feeding coil circuit 1120 (acircuit in which the capacitor C2 and the feeding coil L2 are connectedin series). The horizontal axis represents the drive frequency fo. Theimpedance Z is a minimum value Zmin at the resonance state. AlthoughZmin=0 at the resonance state is ideal, Zmin does not become zero ingeneral since some resistance components are included in the feedingcoil circuit 1120.

When the drive frequency fo and resonance frequency fr1 coincide witheach other, the impedance Z becomes minimum and the capacitor C2 and thefeeding coil L2 are in a resonance state. When the drive frequency foand resonance frequency fr1 deviate from each other, one of thecapacitive reactance and inductive reactance prevails the other, so thatthe impedance Z is also increased.

The impedance Z increases as the deviation from the drive frequency foand resonance frequency fr1 advances, with the result that the powertransmission efficiency is degraded. That is, it is possible to changethe power transmission efficiency by changing the difference between thedrive frequency fo and resonance frequency fr1.

FIG. 19 is a graph illustrating a relationship between the output powerefficiency and drive frequency fo. The output power efficiency is aratio of power actually fed from the feeding coil L2 relative to themaximum output value. When the drive frequency fo coincides with theresonance frequency fr1, a difference between the current phase andvoltage phase becomes zero and therefore the power transmissionefficiency becomes maximum, with the result that output power efficiencyof 100(%) can be obtained. In the wireless power transmission system1100 of the second embodiment, the drive frequency fo is adjusted in arange of fo1 to fo2 which is lower than the resonance frequency fr1.

FIG. 20 is a circuit diagram of the control signal generation circuit1170. The T4 signal (control signal) output from the control signalgeneration circuit 1170 is input to the positive terminal of thecomparator 1174. The T0 signal output from the measurement circuit 1126is input to the negative terminal of the comparator 1174. The T0 signalis a DC voltage signal indicating the correction voltage.

The control signal generation circuit 1170 generates, as the T4 signal,AC voltage of a saw-tooth waveform at the control frequency of 1.0 kHz.The control signal generation circuit 1170 includes resistors R5 to R7,a capacitor C6, and a thyristor 1138. Gate voltage VG obtained bydividing power supply voltage of a power supply VCC by the resistors R5and R6 is applied to the gate G of the thyristor 1138. The gate voltageVG is a fixed value. The anode A of the thyristor 1138 is connected tothe power supply VCC and the ground through the resistor R7 andcapacitor C6, respectively. The power supply voltage is reduced acrossthe resistor R7 and thereby anode potential VA is applied to thethyristor 1138. The T4 signal represents this anode potential VA.

When the anode potential VA is not higher than the gate voltage VG,electric conduction is not provided between the anode and cathode of thethyristor 1138, and the capacitor C6 is charged during this period.After completion of the charging of the capacitor C6, the anodepotential VA (T4 signal) is increased and, when the anode potential VAbecomes higher than the gate potential VG, electric conduction isprovided between the anode and cathode of the thyristor 1138. At thistime, the electrical charge of the capacitor C6 is discharged throughthe thyristor 1138, with the result the anode potential VA becomes nothigher than gate voltage VG once again. The control signal generationcircuit 1170 repeats the above process at 1.0 kHz (control frequencyfc). As a result, the T4 signal of a saw-tooth waveform is generated asdescribed later in FIG. 21. The control frequency fc is determined bythe time constants of the capacitor C6 and resistor R7.

The comparator 1174 outputs a high-level T2 signal (enable signal) whenthe level of the T4 signal is higher than that of the T0 signal while itoutputs a low-level T2 signal in the rest of the time. That is, theperiod during which the level of the T4 signal is higher than that ofthe T0 signal is the enable period, and the rest of the time period isthe disable period. The higher the correction potential, the higher thelevel of the T0 signal, and the shorter the enable period.

FIG. 21 is a time chart illustrating a relationship among the T0, T2, T4signals. In the control signal generation circuit 1170, charging of thecapacitor C6 is started at time t0. The anode potential VA is graduallyincreased and, accordingly, the level of the T4 signal is graduallyincreased. At time t1, the anode potential VA becomes higher than thegate potential VG, and electric conduction is provided between the anodeand cathode of the thyristor 1138. Since the capacitor C6 discharges itselectrical charge, the anode potential VA (T4 signal) is abruptlyreduced. The time period from time t0 to time t1 is referred to as “unitperiod”. The same is applied to the time period after the time t1. Sincethe control frequency fc is 1.0 kHz, the length of each of the unitperiod is 1.0 (msec).

The T0 signal is a DC voltage signal whose voltage level changesdepending on the correction voltage. The comparator 1174 compares the T0signal and T4 signal and generates a high-level T2 signal when the levelof the T4 signal is higher than that of the T0 signal while it outputs alow-level T2 signal in the rest of the time. Among the unit period fromt0 to t1, the T2 signal assumes a low level from time t0 to t4 andassumes a high level from time t4 to t1. That is, among the unit periodfrom time t0 to t1, the time period from time t0 to t4 is the disableperiod, and time period from time t4 to time t1 is enable period. Thelevel of the T0 signal is changed by the correction voltage, causing theduty ratio between the enable period and disable period to change. Whenthe load voltage V5 increases, the correction potential decreases, withthe result that the duty ratio of the T2 signal increases. Conversely,when the load voltage V5 decreases, the correction potential increases,with the result that the duty ratio of the T2 signal decreases. In thepresent embodiment, a setting has been made such that the duty ratiodoes not reach 100% even if the correction potential becomes zero.

FIG. 22 is a circuit diagram of the signal transmission circuit 1122 inthe second embodiment. The signal transmission circuit 1122 includes aninfrared ray LED (Light Emitting Diode) 1158, a transistor Q3, and asignal control circuit 1156. The transistor Q3 is an emitter-groundedbipolar transistor, and the base and emitter thereof are connectedthrough a resistor R9. One end of the infrared ray LED 1158 is connectedto the power supply VCC through the resistor R8, and the other endthereof is connected to the collector of the transistor Q3. The signalcontrol circuit 1156 is also connected to the base of the transistor Q3.

As the signal control circuit 1156, which is a known circuit, an IC(Integrated Circuit) having product serial number UCC37321 manufacturedby Texas Instruments Inc. can be used. The reference signal generationcircuit 1172 and comparator 1174 are connected to the signal controlcircuit 1156. The signal control circuit 1156 receives the T2 and T3signals and outputs a T6 signal. The reference signal generation circuit1172 is an oscillator for generating the T3 signal (reference signal) ata predetermined reference frequency fs. In the second embodiment, thereference frequency fs is assumed to be set to 38 kHz which issufficiently higher than the control frequency fc. Although the waveformof the T3 signal may be a sine wave, it is assumed here that thewaveform of the T3 signal is a rectangular wave (digital waveform).

The signal control circuit 1156 outputs the T3 signal (reference signal)as T6 signal when the T2 signal is high, that is, during the enableperiod. During the disable period, the T6 signal is fixed to a lowlevel.

The T6 signal (AC voltage signal) is changed into a T1 signal (AC lightsignal) by the infrared ray LED 1158. The infrared ray LED 1158transmits the T1 signal (AC light signal) to the signal receivingcircuit 1112. The T1 signal in the second embodiment is an infrared raysignal. The general wavelength of an infrared ray is about 940 nm. TheT1 signal travels up to several meters, so that there occurs no problemeven if the inter-coil distance is large. Further, the infrared ray ishardly subject to the magnetic field generated by the feeding coil L2 orreceiving coil L3, an advantage that the T1 signal and feeding powerhardly interact with each other can be obtained.

FIG. 23 is a time chart illustrating a relationship among the T2, T3,and T6 signals. As described using FIG. 20, the T2 signal (enablesignal) is an AC voltage signal having a control frequency fc of 1.0 kHzin which each of the time periods from t0 to t1, t1 to t2, . . . , isset as the unit period. The time period during which the T2 signalassumes a high level is the enable period, and time period during whichthe T2 signal assumes a low level is the disable period. The T3 signalis an AC voltage signal having a reference frequency fs of 38 kHz. Thesignal control circuit 1156 outputs the T3 signal as the T6 signal onlyduring the enable signal. That is, the logical AND between the T2 and T3signals corresponds to the T6 signal.

The T6 signal which is an AC voltage signal is converted into the T1signal which is the AC light signal and emitted toward the signalreceiving circuit 1112. The infrared ray LED 1158 blinks at thereference frequency fs of 38 kHz during the enable period and turns offin the disable period. The blinking period and turn-off period arerepeated at the control frequency fc. The duty ratio between theblinking period and turn-off period changes depending on the correctionvoltage. The higher the correction voltage is, the shorter the blinkingperiod. The ratio of the blinking period relative to the entire unitperiod is referred to as “duty ratio of T1 signal (output signal)”.

The T2 signal may be used in place of the T6 signal to turn on theinfrared ray LED 1158. In this case, the infrared ray LED 1158 continuesto light during the enable period. The adoption of the configuration ofthe second embodiment in which the infrared ray LED 1158 is made toblink according to the T3 signal during the enable time effectivelyreduces the power consumption of the infrared ray LED 1158.

FIG. 24 is a circuit diagram of the current phase detection circuit 1144and signal receiving circuit 1112 in the second embodiment. The currentphase detection circuit 1144 includes a comparator 1166 and a currentwaveform shaping circuit 1168. The potential VSS is shaped into the S2signal of a digital waveform by the comparator 1142 and input to thecurrent waveform shaping circuit 1168. The current waveform shapingcircuit 1168 shapes the S2 signal of a digital waveform (rectangularwaveform) into an S3 signal of a saw-tooth waveform. In the currentwaveform shaping circuit 1168, a resistor R10 is inserted in the path ofthe S2 signal, and a diode D1 is connected in parallel to the resistorR10. The transmission path of the S2 signal is grounded through acapacitor C7. The S3 signal (AC voltage signal) is input to the positiveterminal of the comparator 1166. The S3 signal is a signal indicatingthe current phase.

The signal receiving circuit 1112 includes a photodiode 1160, a voltageconversion section 1164, and a low-pass filter 1176. The voltageconversion section 1164 includes a comparator 1162 and a resistor R12.

The photodiode 1160 receives the intermittently blinking T1 signal. TheT1 signal (AC light signal) is converted into a T7 signal (AC voltagesignal) by the voltage conversion section 1164. In the voltageconversion section 1164, the resistor R12 is adjusted in the way output1 (my) per 1 lux. The brightness of the T1 signal at the reception timeis about 0 to 2000 (lux) and, accordingly, the voltage level of the T7signal is 0 to 2.0 (V). The duty ratio of the T7 signal indicates thecorrection voltage. The T7 signal (AC voltage signal) is converted intothe T5 signal (DC voltage signal) having a fixed value by the low-passfilter 1176. The low-pass filter 1176 is a general circuit including aresistor R11 and a capacitor C8. The higher the correction voltage, thelower the voltage level of the T5 signal is set. The T5 signal (DCvoltage signal) is input to the negative terminal of the comparator1166. The T5 signal is a DC voltage signal indicating the correctionvoltage.

The comparator 1166 outputs a high-level S1 signal when the level of theS3 signal is higher than that of the T5 signal while it outputs alow-level S1 signal in the rest of the time.

FIG. 25 is a time chart illustrating a relationship among the S1, S3(S2), and T5 signals. The S3 signal is an AC voltage signal of the drivefrequency fo. The S3 signal indicates the current phase. The level ofthe S3 signal starts increasing at time t10 and abruptly decreases attime t11. The time period from time t10 to time t11 corresponds to theunit period of the S3 signal. Since the drive frequency fo is 90 to 99kHz, the time length of the unit period is around 0.01 (msec).

The T5 signal is a DC voltage signal whose voltage level changesdepending on the correction voltage. The comparator 1166 compares the S3signal and T5 signal and generates a high-level S1 signal when the levelof the S3 signal is higher than that of the T5 signal while it outputs alow-level S1 signal in the rest of the time. Among the unit period fromt10 to t11, the S2 signal assumes a low level from time t10 to t14 andassumes a high level from time t14 to t11. The level of the T5 signal ischanged by the correction voltage, causing the duty ratio of the S1signal to change. Although the details will be described later, when theload voltage V5 increases, the correction potential decreases, and thelevel of the T5 signal increases. As a result, the duty ratio of the S1signal decreases, and the rising time of the S1 signal occurs later thanthat of the S3 signal.

FIG. 26 is a time chart illustrating a relationship between the S1signal and S0 signal. Time period from time t20 to time t21(hereinafter, referred to as “first period”) is a time period duringwhich the switching transistor Q1 is ON while the switching transistorQ2 is OFF. Time period from time t21 to time t22 (hereinafter, referredto as “second period”) is a time period during which the switchingtransistor Q1 is OFF while the switching transistor Q2 is ON. Timeperiod from time t22 to time t23 (hereinafter, referred to as “thirdperiod”) is a time period during which the switching transistor Q1 is ONwhile the switching transistor Q2 is OFF. Time period from time t23 totime t24 (hereinafter, referred to as “fourth period”) is a time periodduring which the switching transistor Q1 is OFF while the switchingtransistor Q2 is ON.

At time t20, the AC voltage Vo (S0 signal) changes from the minimumvalue to the maximum value. At time t21 at which the first time periodis ended, the AC voltage Vo (S0 signal) changes from the maximum valueto the minimum value. Hereinafter, a timing (represented by, e.g., timet20) at which the S0 signal rises is referred to as “voltage phasevalue”.

In the case where the drive frequency fo is lower than the resonancefrequency fr1, a capacitive reactance component appears in the impedanceZ of the feeding coil circuit 1120 (LC resonance circuit), and thecurrent phase of the current Is advances with respect to the voltagephase. Thus, the S2 signal indicating a current phase rises at time t10which is earlier than time t20. Hereinafter, a timing (represented by,e.g., time t10 at which the S2 signal rises is referred to as “currentphase value”. In the example of FIG. 26, a value obtained by t20-t10represents the phase difference. Here, t20-t10>0 is established, so thatthe current phase advances with respect to the voltage phase.

When the S2 signal rises at time t10, the level of the S3 signal startsincreasing. At time t11 at which the level of the S2 signal becomeszero, the level of the S3 signal also abruptly decreases.

The T5 signal is a DC voltage signal whose level changes depending onthe magnitude of the correction voltage. In FIG. 26, the correctionvoltage is detected, that is, the load voltage V5 deviates from adesired value.

The S3 signal and T5 signal are input to the positive terminal andnegative terminals of the comparator 1166, respectively, and the S1signal is output from the comparator 1166. During the level of S3 signalis higher than that of T5 signal, the level of the S1 signal is sethigh, while in the rest of time, the level of the S1 signal is set low.In FIG. 26, the level of the S3 signal is higher than that of the T5(S3>T5) at time t14 (hereinafter, such a timing is referred to also as“corrected current phase value”) which is later than time t10. Thevoltage level of the T5 signal serves as a “reference value” fordetermining the corrected current phase value.

The phase comparison circuit 1150 compares rising edge time t10 of theS0 signal and rising edge time t14 of the S1 signal to calculate thephase difference td. Although the actual phase difference is t20-t10(>0), the phase difference recognized by the phase comparison circuit1150 is obtained by t20-t14 (<0). The phase comparison circuit 1150outputs the phase difference indicating voltage SC corresponding to avalue obtained by t20-t14. The VCO 1202 determines that the currentphase delays with respect to the voltage phase, that is, the drivefrequency fo is higher than the resonance frequency fr1 and tries toeliminate the phase difference by reducing the drive frequency fo. As aresult, feedback control is executed such that the power transmissionefficiency is degraded, the load voltage V5 is reduced, and thecorrection voltage is eliminated.

For example, the resistance value of the load LD increases, the loadcurrent I5 decreases while the load voltage V5 increases (refer to FIG.16). When the load voltage V5 increases, the measurement potentialincreases while the correction voltage decreases. As a result, thevoltage level of the T0 signal (DC voltage signal) decrease.

When the voltage level of the T0 signal decreases, the duty ratio of theT2 signal increases (refer to FIG. 21). As a result, the duty ratio ofthe T1 signal (output signal) also increases (refer to FIG. 23). Whenthe duty ratio of the T1 signal increases, the voltage level of the T5signal (DC voltage signal) increases. As a result, the duty ratio of theS1 signal decreases. Further, the rising edge of the S1 signal occurslater than that of the S0 signal, so that the phase comparison circuit1150 recognizes that current phase delays with respect to the voltagephase. In order to eliminate the delay of the current phase, the phasecomparison circuit 1150 issues the phase difference indicating voltageSC to the VCO 1202 for decreasing the drive frequency fo. Then, thedeviation between the resonance frequency fr1 and drive frequency fobecomes larger to cause the power transmission efficiency to decrease(refer to FIGS. 18 and 19), with the result that the load voltage V5decreases. With such feedback control, the load voltage V5 can be keptat a fixed value. The same feedback control is performed when the loadvoltage V5 decreases.

FIG. 27 is a circuit diagram illustrating a modification example of thewireless power receiver 1118 in the second embodiment. Although the DCcurrent I5 is fed to the load LD in FIG. 15, the AC current I4 maydirectly be fed to the load LD in the modification example. In thiscase, the rectification circuit 1124 and measurement circuit 1126 areconnected to a part of the load coil L4 so as to allow the T0 signal tobe output.

FIG. 28 is a system configuration view of a wireless power transmissionsystem 1100 which is a modification of the second embodiment. In thewireless power transmission system 1100 of the modification, the powertransmission control circuit 1200 directly drives the feeding coilcircuit 1120 without intervention of the exciting circuit 1110.Components designated by the same reference numerals as those of FIG. 2have the same or corresponding functions as those in FIG. 2.

The feeding coil circuit 1120 in the modification is a circuit in whichthe transformer TS2 secondary coil Li is connected in series to thefeeding coil L2 and capacitor C2. The transformer TS2 secondary coil Liconstitutes a coupling transformer TS2 together with the transformer TS2primary coil Lb and receives AC power from the power transmissioncontrol circuit 1200 by electromagnetic induction. Thus, the AC powermay be directly fed from the power transmission control circuit 1200 tothe feeding coil circuit 1120 without intervention of the excitingcircuit 1110.

FIG. 29 is a view illustrating operation principle of the wireless powertransmission system 100 according to a third and fourth embodiments. Asin the case of the first and second embodiments, the wireless powertransmission system 100 according to the third and fourth embodimentsincludes the wireless power feeder 116 and wireless power receiver 118.However, although the wireless power receiver 118 includes the powerreceiving LC resonance circuit 302, the wireless power feeder 116 doesnot include the power feeding LC resonance circuit 300. That is, thefeeding coil L2 does not constitute a part of the LC resonance circuit.More specifically, the feeding coil L2 does not form any resonancecircuit with other circuit elements included in the wireless powerfeeder 116. No capacitor is connected in series or in parallel to thefeeding coil L2. Thus, the feeding coil L2 does not resonate in afrequency at which power transmission is performed.

The power feeding source VG supplies AC current of the resonancefrequency fr1 to the feeding coil L2. The feeding coil L2 does notresonate but generates an AC magnetic field of the resonance frequencyfr1. The receiving LC resonance circuit 302 resonates by receiving theAC magnetic field. As a result, large AC current flows in the powerreceiving LC resonance circuit 302. Studies conducted by the presentinventor have revealed that formation of the LC resonance circuit is notessential in the wireless power feeder 116. The feeding coil L2 does notconstitute a part of the power feeding LC resonance circuit, so that thewireless power feeder 116 does not resonate at the resonance frequencyfr1. It has been generally believed that, in the wireless power feedingof a magnetic field resonance type, making resonance circuits which areformed on both the power feeding side and power receiving side resonateat the same resonance frequency fr1 (=fr0) allows power feeding of largepower. However, it is found that even in the case where the wirelesspower feeder 116 does not contain the power feeding LC resonance circuit300, if the wireless power receiver 118 includes the power receiving LCresonance circuit 302, the wireless power feeding of a magnetic fieldresonance type can be achieved.

Even when the feeding coil L2 and receiving coil L3 aremagnetic-field-coupled to each other, a new resonance circuit (newresonance circuit formed by coupling of resonance circuits) is notformed due to absence of the capacitor C2. In this case, the strongerthe magnetic field coupling between the feeding coil L2 and receivingcoil L3, the greater the influence exerted on the resonance frequency ofthe power receiving LC resonance circuit 302. By supplying AC current ofthis resonance frequency, that is, a frequency near the resonancefrequency fr1 to the feeding coil L2, the wireless power feeding of amagnetic field resonance type can be achieved. In this configuration,the capacitor C2 need not be provided, which is advantageous in terms ofsize and cost.

Third Embodiment

FIG. 30 is a system configuration view of the wireless powertransmission system 100 according to the third embodiment. In thewireless power transmission system 100 of the third embodiment, thecapacitor C2 is omitted. Other points are the same as the firstembodiment (FIG. 14).

Fourth Embodiment

FIG. 31 is a system configuration view of the wireless powertransmission system 100 according to the fourth embodiment. In thewireless power transmission system 100 of the fourth embodiment, thecapacitor C2 is omitted. Other points are the same as the secondembodiment (FIG. 28).

The wireless power transmission system has been described above based onthe embodiments. In the wireless power feeding of a magnetic fieldresonance type, the power transmission efficiency can be controlled bythe difference between the resonance frequency fr1 and drive frequencyfo. The drive frequency fo can be made to automatically track a changeof the resonance frequency fr1, making it easy to kept the powertransmission efficiency constant even if use conditions are changed.Further, even when the load LD or inter-coil distance d is changed, theload voltage V5 can be kept constant by the feedback control based onthe correction voltage. A change of the level of the S1 signal based onthe correction voltage allows ex-post adjustment of the powertransmission efficiency. According to the experiment made by the presentinventor, significant power loss was not found to occur in associationwith the level adjustment of the S1 signal.

The T0 signal which is a DC voltage signal is converted into the T1signal which is an AC light signal, and the T1 signal is emitted fromthe wireless power receiver to the wireless power feeder. The T1 signalas the light signal is hardly subject to the magnetic field generated bythe feeding coil L2 and the like, an advantage that signals canfavorably be transmitted can be obtained.

Further, the reference potential may manually be adjusted on the powerreceiving side. This allows the correction voltage to be detected notonly when the measurement potential is changed but also the referencevoltage is changed, with the result that the power transmissionefficiency can be adjusted. For example, when the reference potential ismade to decrease, such feedback control as to decrease the measurementpotential is made, casing the load voltage V5 to decrease. That is,feeding power can be controlled on the power receiving side.

As an application example, the following configuration may be possible.That is, the wireless power feeder and a table are integrated with eachother, and the wireless receiver is incorporated in a table lamp placedon the table. In the case of a conventional table lamp, a power cordgets in the way, so that a pendant lamp is often used for a diningtable. According to the above application example, it is possible toeliminate the need of providing a power cord of the table lamp, therebyincreasing availability of the table lamp. For example, there may be acase where food looks more attractive under illumination of the tablelamp. Further, the illumination location is fixed in the case of thependant lamp, while the table lamp can freely be laid out, enablingvarious forms of illumination. In addition, in the case where aplurality of table lamps are placed on the table, adjusting thereference potential of only one table lamp can collectively control thebrightness of other lamps.

The above embodiments are merely illustrative of the present inventionand it will be appreciated by those skilled in the art that variousmodifications may be made to the components of the present invention anda combination of processing processes and that the modifications areincluded in the present invention.

Although the power transmission control circuit 1200 is formed as ahalf-bridge type circuit in the above embodiment, the power transmissioncontrol circuit 1200 may be formed as a push-pull type circuit orfull-bridge type circuit. The S3 signal generated by the currentwaveform shaping circuit 1168 or T4 signal generated by the controlsignal generation circuit 1170 may be an AC signal having not only asaw-tooth waveform but also a triangle wave or a sine wave in which avoltage value is gradually increased or decreased within a predeterminedtime period. Although the current phase is set as an adjustment targetin the present embodiment, the voltage phase may be adjusted based onthe T0 signal. Further, the feedback control may be effected based notonly on the output voltage but on the current or power.

The T1 signal is not limited to a light signal such as an infrared raybut may be a radio signal. At any rate, it is only necessary for the T1signal to have a frequency band sufficiently away from the frequencyband of the drive frequency fo or resonance frequency fr1. The infraredray LED 1158 and photodiode 1160 are comparatively low in price, so thatthe light signal is adopted in the present embodiments.

The “AC power” used in the wireless power transmission system 100 may betransmitted not only as an energy but also as a signal. Even in the casewhere an analog signal or digital signal is fed by wireless, thewireless power transmission method of the present invention may be used.

What is claimed is:
 1. A wireless power feeder for feeding power from afeeding coil to a receiving coil by wireless using a magnetic fieldresonance phenomenon between the feeding coil and receiving coil: afeeding coil circuit that includes the feeding coil; a powertransmission control circuit that feeds AC power to the feeding coil ata drive frequency; a phase detection circuit that detects a phasedifference between the voltage phase and current phase of the AC power;and a signal receiving circuit that receives an output signal indicatingthe magnitude of an output from the power receiving side of the ACpower, wherein the power transmission control circuit adjusts the drivefrequency so as to reduce the phase difference, and the phase detectioncircuit performs ex-post adjustment of the detected value of both or oneof the voltage and current phases according to the output signal.
 2. Thewireless power feeder according to claim 1, wherein the phase detectioncircuit converts both or one of voltage and current components of the ACpower into a signal having a saw-tooth waveform for detection of thephase difference.
 3. The wireless power feeder according to claim 1,wherein the signal receiving circuit receives the output signal as alight signal.
 4. The wireless power feeder according to claim 1, whereinthe phase detection circuit compares a first phase value indicating atiming at which the voltage level of the AC power becomes a firstreference value and a second phase value indicating a timing at whichthe current level of the AC power becomes a second reference value todetect the phase difference and changes both or one of the first andsecond reference values based on the output signal to perform ex-postadjustment of both or one of the first and second phase values.
 5. Thewireless power feeder according to claim 1, wherein the output signal isan AC signal indicating the magnitude of the output by the magnitude ofsignal frequency.
 6. The wireless power feeder according to claim 1,further comprising an exciting coil that is magnetically coupled to thefeeding coil and feeds AC power fed from the power transmission controlcircuit to the feeding coil, wherein the power transmission controlcircuit includes first and second current paths and makes first andsecond switches connected in series respectively to the first and secondcurrent paths alternately conductive at the drive frequency to feed theAC power to the exciting coil.
 7. The wireless power feeder according toclaim 1, further comprising a detection coil that generates inductivecurrent using a magnetic field generated by the AC power, wherein thephase detection circuit measures the phase of the inductive currentflowing in the detection coil to achieve measurement of the currentphase of the AC power.
 8. The wireless power feeder according to claim1, wherein the power supply circuit makes the feeding coil that does notsubstantially resonate with circuit elements on the power feeding sidefeed the AC power to the receiving coil.
 9. The wireless power feederaccording to claim 1, wherein the feeding coil does not form, togetherwith circuit elements on the power feeding side, a resonance circuithaving a resonance point corresponding to the resonance frequency of thereceiving coil.
 10. The wireless power feeder according to claim 1,wherein no capacitor is connected in series or in parallel to thefeeding coil.
 11. The wireless power feeder according to claim 1,wherein the feeding coil circuit resonates at the resonance frequency ofthe receiving coil.
 12. A wireless power receiver that receives AC powerfed from the wireless power feeder as claimed in claim 1 by wireless ata receiving coil, the receiver comprising: a receiving coil circuit thatincludes the receiving coil and a capacitor; a loading circuit thatincludes a loading coil that is magnetically coupled to the receivingcoil to receive the AC power from the receiving coil and a load thatreceives power from the loading coil; and a signal transmission circuitthat transmits, to the wireless power feeder, an output signalindicating the magnitude of output voltage to be applied to a part ofthe loading circuit.
 13. The wireless power receiver according to claim12, wherein the signal transmission circuit transmits the output signalas a signal indicating a difference value between the output voltage anda reference voltage.
 14. The wireless power receiver according to claim13, wherein the value of the reference voltage is adjustable.
 15. Thewireless power receiver according to claim 12, wherein the output signalis an AC signal indicating the magnitude of the output voltage by themagnitude of signal frequency.
 16. The wireless power receiver accordingto claim 12, wherein the loading circuit includes a rectificationcircuit, and the output voltage is generated by the rectificationcircuit as a DC current.
 17. The wireless power receiver according toclaim 12, wherein the receiving coil circuit resonates at the resonancefrequency of the feeding coil.
 18. A wireless power transmission systemfor feeding power from a feeding coil to a receiving coil by wirelessusing a magnetic field resonance phenomenon between the feeding coil andreceiving coil, comprising: a feeding coil circuit that includes thefeeding coil; a power transmission control circuit that feeds AC powerto the feeding coil at a drive frequency; a receiving coil circuit thatincludes the receiving coil and a capacitor; a loading circuit thatincludes a loading coil that is magnetically coupled to the receivingcoil to receive the AC power from the receiving coil and a load thatreceives power from the loading coil; and a phase detection circuit thatdetects a phase difference between the voltage phase and current phaseof the AC power, wherein the power transmission control circuit adjuststhe drive frequency so as to reduce the phase difference, and the phasedetection circuit performs ex-post adjustment of the detected value ofboth or one of the voltage and current phases according to the magnitudeof an output voltage to be applied to a part of the loading circuit. 19.A wireless power feeder for feeding power from a feeding coil to areceiving coil by wireless using a magnetic field resonance phenomenonbetween the feeding coil and receiving coil, comprising: a feeding coilcircuit that includes the feeding coil; a power transmission controlcircuit that feeds AC power to the feeding coil at a drive frequency; aphase detection circuit that detects a phase difference between thevoltage phase and current phase of the AC power; and a signal receivingcircuit that receives an output signal indicating an output by a dutyratio from the power receiving side of the AC power and DC-converts theoutput signal in accordance with the duty ratio, wherein the powertransmission control circuit adjusts the drive frequency so as to reducethe phase difference, and the phase detection circuit performs ex-postadjustment of the detected value of both or one of the voltage andcurrent phases in accordance with the signal level of the DC-convertedoutput signal.
 20. The wireless power feeder according to claim 19,wherein the phase detection circuit compares a first phase valueindicating a timing at which the voltage level of the AC power becomes afirst reference value and a second phase value indicating a timing atwhich the current level of the AC power becomes a second reference valueto detect the phase difference and changes both or one of the first andsecond reference values in accordance with the signal level to performex-post adjustment of both or one of the first and second phase values.21. The wireless power feeder according to claim 19, wherein the signalreceiving circuit receives the output signal as a light signal.
 22. Thewireless power feeder according to claim 19, further comprising anexciting coil that is magnetically coupled to the feeding coil and feedsAC power fed from the power transmission control circuit to the feedingcoil, wherein the power transmission control circuit includes first andsecond current paths and makes first and second switches connected inseries respectively to the first and second current paths alternatelyconductive at the drive frequency to feed the AC power to the excitingcoil.
 23. The wireless power feeder according to claim 19, furthercomprising a detection coil that generates inductive current using amagnetic field generated by the AC power, wherein the phase detectioncircuit measures the phase of the inductive current flowing in thedetection coil to achieve measurement of the current phase of the ACpower.
 24. The wireless power feeder according to claim 19, wherein thepower supply circuit makes the feeding coil that does not substantiallyresonate with circuit elements on the power feeding side feed the ACpower to the receiving coil.
 25. The wireless power feeder according toclaim 19, wherein the feeding coil does not form, together with circuitelements on the power feeding side, a resonance circuit having aresonance point corresponding to the resonance frequency of thereceiving coil.
 26. The wireless power feeder according to claim 19,wherein no capacitor is connected in series or in parallel to thefeeding coil.
 27. The wireless power feeder according to claim 19,wherein the feeding coil circuit resonates at the resonance frequency ofthe receiving coil.
 28. A wireless power receiver that receives AC powerfed from the wireless power feeder as claimed in claim 19 by wireless ata receiving coil, the receiver comprising: a receiving coil circuit thatincludes the receiving coil and a capacitor; a loading circuit thatincludes a loading coil that is magnetically coupled to the receivingcoil to receive the AC power from the receiving coil and a load thatreceives power from the loading coil; and a signal transmission circuitthat transmits, to the wireless power feeder, an output signalindicating output voltage to be applied to a part of the loading circuitby a duty ratio.
 29. The wireless power receiver according to claim 28,wherein the signal transmission circuit transmits the output signal as asignal indicating a difference value between the output voltage and areference voltage by the duty ratio.
 30. The wireless power receiveraccording to claim 29, wherein the value of the reference voltage isadjustable.
 31. The wireless power receiver according to claim 28,further comprising: a control signal generation circuit that generates acontrol signal at a predetermined control frequency; and a comparisoncircuit that generates an enable signal when a predetermined magnituderelationship is established between the signal level of the controlsignal and the output voltage, wherein the signal transmission circuitdetermines the duty ratio of the output signal based on the duty ratioof the enable signal.
 32. The wireless power receiver according to claim31, further comprising a reference signal generation circuit thatgenerates a reference signal having a reference frequency higher thanthe control frequency, wherein the signal transmission circuit transmitsthe reference signal as the output signal only while the enable signalis being activated.
 33. The wireless power receiver according to claim28, wherein the receiving coil circuit resonates at the resonancefrequency of the feeding coil.
 34. A wireless power transmission systemfor feeding power from a feeding coil to a receiving coil by wirelessusing a magnetic field resonance phenomenon between the feeding coil andreceiving coil, comprising: a feeding coil circuit that includes thefeeding coil; a power transmission control circuit that feeds AC powerto the feeding coil at a drive frequency; a receiving coil circuit thatincludes the receiving coil and a capacitor; a loading circuit thatincludes a loading coil that is magnetically coupled to the receivingcoil to receive the AC power from the receiving coil and a load thatreceives power from the loading coil; a phase detection circuit thatdetects a phase difference between the voltage phase and current phaseof the AC power; a signal transmission circuit that transmits, to thepower feeder side, an output signal indicating output voltage to beapplied to a part of the loading circuit by a duty ratio; and a signalreceiving circuit that receives the output signal at the power feederside and DC-converts the output signal in accordance with the dutyratio, wherein the power transmission control circuit adjusts the drivefrequency so as to reduce the phase difference, and the phase detectioncircuit performs ex-post adjustment of the detected value of both or oneof the voltage and current phases in accordance with the signal level ofthe DC-converted output signal.